A Millimeter-Wave Antenna With Filtering Radiation Response Based on Absorptive Ring Feeding Circuit

Abstract

This letter presents and analyzes an absorptive filtering antenna design operating at 5G millimeter-wave (mm-wave) band. It consists of a two-port ring feeding circuit, a magneto-electric dipole (ME-dipole), and a square metal loop. The two-port ring circuit integrates bandstop-type transmission performance. Within the stopband of this two-port ring circuit, two controllable transmission zeros are deliberately generated to prevent the input signal transmitting from one port to the other. As a result, the antenna operates as an effective radiator within this band. However, at the upper band frequencies, the input signal transmits through one port to the other and thus the antenna does not work efficiently. To further improve the out-of-band radiation suppression level, a square metal loop around the ME-dipole arms is used. Based on such a simple structure, bandpass-type filtering radiation response is realized. For demonstration, a filtering antenna array for 5G mm-wave applications (n257/n258/n261 bands: 24.25∼29.5 GHz) is implemented and tested as an example. The measured average in-band gain is about 11.5 dBi and better than 16 dB out-of-band radiation rejection level is achieved.

Full text

1
1
Abstract—This letter presents and analyzes an absorptive
filtering antenna design operating at 5G millimeter-wave
(mm-wave) band. It consists of a two-port ring feeding circuit, a
magneto-electric dipole (ME-dipole), and a square metal loop.
The two-port ring circuit integrates bandstop-type transmission
performance. Within the stopband of this two-port ring circuit,
two controllable transmission zeros are deliberately generated to
prevent the input signal transmitting from one port to the other.
As a result, the antenna operates as an effective radiator within
this band. However, at the upper band frequencies, the input
signal transmits through one port to the other and thus the
antenna does not work efficiently. To further improve the
out-of-band radiation suppression level, a square metal loop
around the ME-dipole arms is used. Based on such a simple
structure, bandpass-type filtering radiation response is realized.
For demonstration, a filtering antenna array for 5G mm-wave
applications (n257/n258/n261 bands: 24.25~29.5 GHz) is
implemented and tested as an example. The measured average
in-band gain is about 11.5 dBi and better than 16 dB out-of-band
radiation rejection level is achieved.
Index Terms—Filtering antenna, millimeter-wave, bandpass
filter, magneto-electric dipole, filtering radiation response.
I. INTRODUCTION
illimeter-wave antenna has received much attention in
the past decade due to its significance to communication
systems with high data rate and wide operating bandwidth [1].
For traditionally mm-wave antenna-in-package (AIP) designs,
filters and antennas operate separately which cause increased
system insertion loss and extra chip size. The filtering antenna
technique has potential value to solve this problem [2].
Over the past few years, the filtering antenna technique has
drawn great attention for its contribution of mutual coupling
reduction in sub-6 GHz multi-band base-station array antenna
designs [3]-[6]. By replacing the last order resonator of a
band-pass filter with the antenna radiator, some filtering
1
Manuscript received Apr. 26, 2023, revised Jun. 20, 2023.
This work was supported in part by the National Natural Science Foundation
of China under Grant 62001407 and Grant U2241222 (Corresponding author:
Yao Zhang).
K. Huang and L. F. Ye are with the Institute of Electromagnetics and
Acoustics, Xiamen University, Xiamen, 361005, China.
Y. Zhang is with the Institute of Electromagnetics and Acoustics, Xiamen
University, Xiamen, 361005, China, also with the Guangdong Provincial Key
Laboratory of Optoelectronic Information Processing Chips and Systems,
Guangzhou 510006, China, and also with the State Key Laboratory of
Millimeter Waves, Nanjing 210096, China (zhangsantu123@sina.cn).
Q. H. Liu is with Eastern Institute of Technology, Ningbo 315100, China,
and also with the Institute of Electromagnetics and Acoustics, Xiamen
University, Xiamen 361005, China.
antennas have been developed in [7]-[9]. In these designs, the
filtering radiation performance was realized but the insertion
loss of the resonator filters still existed. Specific parasitic
components such as slot [10], parasitic patch [11], shorting pins
[12], microstrip stub [13] were employed in antenna structures
to achieve filtering radiation response by generating specific
out-of-band radiation zeros. In this way, without using complex
filtering circuits, more compact size and lower insertion loss
were obtained.
Very recently, an interesting design concept of absorptive
filtering antennas has been proposed [14]-[16]. By loading the
absorptive components such as the branch or resistance, the
antennas obtained a wide reflection-less band based on very
simple structures.
Inspired by this design concept, this letter presents a different
absorptive filtering antenna realization method based on a
two-port ring feeding circuit. This circuit is functioned as not
only a feeding network but also a high-pass absorptive filter.
The stop frequency band is deliberately tuned to the 5G
millimeter-wave band 24.25~29.5 GHz. Within this band, it
ensures a good antenna impedance matching. Whereas at
upper-band frequencies (> 32 GHz), it enables the input signal
transmitted from one port to the other and thus a reflection-less
band is realized. As a result, the antenna does not work at the
upper-band frequencies and radiation suppression is obtained.
II. ANTENNA CONFIGURATION
Fig. 1 depicts the configuration of the proposed absorptive
filtering antenna. It has three layers of substrates denoted as
Sub 1, 2, 3, which are used for the ME-dipole part, the bonding
film and the feeding network. The electric dipole (E-dipole)
arms with length L2 are printed on the top surface of the Sub 1
A Millimeter-Wave Antenna with Filtering Radiation
Response Based on Absorptive Ring Feeding Circuit
Kai Huang, Yao Zhang, Member, IEEE, Longfang Ye, and Qing Huo Liu, Fellow, IEEE
M
Sub1
Sub2
Sub3
Port1 Port2
z
x
y
L8
L7
L6
L5
W50
W2
squareness-loop
cross-shaped
driven patch
L1
G
L2
S
R1
L3
R2
W1L4S1
ring-slot
R3
blind hole
through hole
Fig. 1. Configuration of the proposed antenna element (L1 = 5, L2 = 1.89, L3 =
0.63, R1 = 0.21, S = 0.68, W1 = 0.48, L4 = 2.18, R2 = 0.14, S1 = 0.17, L5 = 2, L6 =
1.7, W2 = 0.5, W3 = 0.16, L7 = 1.7, L8 = 3.35, R3 = 0.375, G = 15, all in mm).
This article has been accepted for publication in IEEE Antennas and Wireless Propagation Letters. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/LAWP.2023.3296150
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: Xiamen University. Downloaded on July 19,2023 at 01:27:34 UTC from IEEE Xplore. Restrictions apply.

2
while the magnetic dipole (M-dipole) is implemented with the
use of a shorted patch antenna which consists of 24 blind holes
with radius R1 and the portion of the ground plane connecting
them together. The blind holes are connected with the E-dipole
arms. A square loop is loaded outside the E-dipole arms. A
cross-shaped driven patch with four through holes is seen,
which couples the input signal from the feedline to the
ME-dipole radiator. Four ring-slot with radius R3 are loaded in
the ground plane for the connection of the through holes. The
Sub1 used for this part is Rogers 5880 with a thickness (h1) of
1.575 mm. The feeding part is a three-dimensional (3-D)
dual-ring configuration. The Sub3 used for the feeding part is
Rogers 3003 with a thickness (h3) of 0.127 mm. Between the
Sub 1 and Sub 3, a bonding film (Rogers 4450F) with a
thickness (h2) of 0.1 mm is used to align them together. Based
on such a configuration, a millimeter-wave antenna with
filtering radiation response is realized. The design procedure
and working principle of this antenna has been studied in detail
in the following sections.
III. ANTENNA EVOLUTION AND MECHANISM
In order to better disclose the evolution of the proposed
absorptive filtering antenna, five related reference antennas
denoted as the original Ant, Ant. I, Ant. II, Ant. III and Ant. IV
are proposed and investigated one by one below.
A. Original Patch Ant. and Reference Patch Ant. I
The design of the proposed filtering antenna is originated
from a single port patch antenna, as seen in Fig. 2. It consists of
a rectangle patch with a rectangle slot and a pair of ideal
differential ports. Based on this structure, a two-port patch
antenna was designed as the Ant. I. It is a traditional E-dipole
antenna consists of four rectangular patches and a
differential-fed cross-shaped driven circuit.
B. Evolution of the Reference Ant. II
The Ant.  is then modified to the Ant. II, as seen in Fig. 2.
Twenty-four metal pins are added to connect the radiation
patches to the ground. Fig. 3 shows the S-parameters and
realized gains. It is noted that the Ant.  has a flat gain of about
8.6 dBi, and it decreases slowly when the frequency deviates
from the operating frequency. However, the Ant. II has a sharp
roll-off-rate at the lower passband edge and a specific radiation
Null 1 is generated at 19.1 GHz.
Deduced from Fig. 3, the radiation Null 1 is generated by
adding the 24 vertical pins. In order to show the effect of the
them, Fig. 4 shows the surface current distribution of the Ant. I
and the Ant. II. As seen, similar uniform currents are seen on
the E-dipole arms for both two antennas. For the Ant. II, it’s
seen that the currents on the vertical pins flow in opposite
directions. The vertical pins here work as the Balun structure
and the radiations of the vertical arms are cancelled out at
boresight direction. Now consider the ME-dipole arm as a
lossless open-circuited transmission line [17]-[18]. If l = (h1 +
L2) =

/4, where

is the wavelength of the specific frequency,
the input impedance is Zin(in case l=λ/4)=0 [19]. In this case, the
point A is equal to be shorted. That means the input signal
cannot pass through this transmission line. Therefore, this type
ME-dipole antenna has an inherent radiation Null 1.
C. Evolution of the Reference Ant. III
The Ant. II has a highpass-type radiation response. To realize
bandpass-type filtering performance, the Ant. III is proposed by
adding a dual-ring feeding circuit. This dual-ring circuit can be
used to suppress the upper out-of-band antenna radiation.
Obviously, the gain of the Ant. III decreases sharply at upper
out-of-band frequencies, as seen in Fig. 3.
In this design, there are two requirements for this feeding
circuit. On the one hand, it should function as an antenna
feeding network with a high reflection level within the antenna
operating band. On the other hand, at upper out-of-band
frequencies, it should enable the input signal transmitted
through one port to the other port to realize an upper
reflection-less band with high suppression level.
To fulfill this requirement, a two-port dual-ring feeding
circuit is developed as shown in Fig. 5. It is a three-dimensional
structure. The transmission paths of the Port 1 and Port 2 are
marked by the green and yellow line, respectively. These two
paths are connected by a cross-shaped driven patch.
To study the working principle of this equivalent
second-order ring feeding circuit, a lossless first-order ring
microstrip line model is investigated first. Its equivalent circuit
is seen in Fig. 6(a). The lengths of the two paths from Port 1 to
Port 2 are denoted as L1 and L2, respectively. Corresponding
S-parameters of this circuit can be calculated by [19]:
patch
patch ME-dipole
Dual-ring circuit Square loop
Original Ant. Ant. I Ant. II
Ant. III Ant. IV
Fig. 2. Evolution of the proposed antenna.
20 25 30 35 40
-30
-25
-20
-15
-10
-5
0
Ant. I
Ant. II
Ant. III
Ant. IV
S-Parameters (dB)
Frequency (GHz)
20 25 30 35 40
-25
-20
-15
-10
-5
0
5
10
Realized Gain (dBi)
Frequency (GHz)
Ant. I
Ant. II
Ant. III
Ant. IV
Null 1 Null 2
Fig. 3. Reflection coefficients and realized gains of the Ant. I, II, III and IV.
Zin A
2.0E+001
8.0E-002
Jsurf [A/m]
Ant. I Ant. II
L2h1
L2
(a) (b)
Fig. 4. Current distributions at 28 GHz of (a) Ant. I and (b) Ant. II.
This article has been accepted for publication in IEEE Antennas and Wireless Propagation Letters. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/LAWP.2023.3296150
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: Xiamen University. Downloaded on July 19,2023 at 01:27:34 UTC from IEEE Xplore. Restrictions apply.

3
12
21 22
12
1 2 1 2
11
2 ( )
sin sin
cos cos 11
( 1) ( )
sin sin sin sin
jLL
SLL
j L j L j L j L


   


   
(1)
where

is the phase constant. From the equation (1), we can
obtain the frequency of the transmission zero under the
condition of S21 = 0, which can be simplified as sinβL1 = -sinβL2.
Then we can get the results L1 + L2 = nλ or L1 - L2 = (2n-1)λ/2.
Where n = 1, 2, 3,… That means once the total length of the
ring circuit is equal to a wavelength of a specific frequency, or
the length difference between the two paths is equal to its
half-wavelength, a transmission zero is occurred. Also, it can
be concluded that when the length (L1 + L2) is fixed to be λ1, a
corresponding transmission zero is obtained. Furthermore,
when the length difference (L2 - L1) = 0.5 λ2, another
transmission zero is generated. Above results reveal that the
ring feeding circuit has two controlled transmission zeros.
Therefore, we firstly select the bandstop frequency of the
feeding circuit to be 28 GHz, that is (L1 + L2) =λ1. Fig. 6(b)
shows the transmission coefficient S12 of the ring feeding
circuit when (L1 + L2) = λ1 and (L2 - L1) = 4.1 mm. In this case,
only one transmission zero at 28 GHz is observed. Secondly,
keep the length (L1 + L2) = λ1 unchanged, by tuning the length
difference (L2 - L1), another transmission zero can be realized
and controlled, as plotted in Fig. 6(c).
Then the first-order ring feeding circuit is modified to a
second-order ring circuit, as shown in Fig. 6(d). On the one
hand, the proposed feeding circuit ensures a good antenna
impedance matching within the antenna band. On the other
hand, it enables the input signal transmitted from one port to the
other port at absorptive band with filtering response.
D. Evolution of the Proposed Ant. IV
To further improve the upper out-of-band radiation
suppression level, a square loop is printed for generating an
upper-band radiation null. As observed in Fig. 3, another
radiation Null 2 is generated at 36.1 GHz, ensuring high upper
out-of-band radiation rejection level.
To investigate the radiation Null 2, the current distribution
on the loop and radiation patch of the proposed Ant. IV at 36.1
GHz is shown in Fig. 7. It can be seen that the current mainly
concentrates on the square-loop and the currents on the antenna
patches are weak. By tuning the size of the square-loop, the
frequency of Null 2 can be individually controlled, and it can be
estimated by the following equation:
22
null
r
c
fL


(2)
where c is the speed of light, 𝜀𝑟 represents the effective
dielectric constant of the substrate, and L is the length of the
1/2-cycle current route. Therefore, a bandpass-type response is
achieved. The frequencies of the two radiation Nulls 1 and 2
also can be controlled individually, as seen in Fig. 8. Also, the
upper cut-off frequency can be manipulated, as seen in Fig. 9(a).
Fig. 9(b) shows the radiation efficiency. The radiation
efficiency in the antenna band is higher than 84%, while that in
the out-of-band are less than 2%.
IV. ANTENNA ARRAY IMPLEMENTATION
Using the proposed absorptive filtering antenna element, a 2
× 2 antenna array is implemented, as depicted in Fig. 10. For
rotate
rotate
Port 1
Port 2 Cross part
Port 1
Port 2
Fig. 5. Structure and equivalent planar circuit of the two-port feeding circuit.
Y2
Y1
Port 1 Port 2
Port 1
Port 2
L1
L2
Port 1
Port 2
L1
L2
20 25 30 35 40
-40
-30
-20
-10
0
Transmission coefficient (dB)
Frequency (GHz)
L1 + L2 = λ1 = 8.2 mm
L2 - L1 = 0.5 λ1
(a) (b)
20 25 30 35 40
-50
-40
-30
-20
-10
0
L1 = 1.6 mm L1 = 1.4 mm
L1 = 1.2 mm
Transmission coefficient (dB)
Frequency (GHz)
L1 + L2 = λ1 = 8.2 mm
L2 - L1 ≠ 0.5 λ1
20 25 30 35 40
-50
-40
-30
-20
-10
0
Port 1
Port 2
Absortive band
S12
Transmission coefficient (dB)
Frequency (GHz)
Antenna band
(c) (d)
Fig. 6. (a) Equivalent circuit, (b)-(c) simulation results of the first-order ring
microstrip line model and (d) simulation transmission coefficients of the
dual-ring resonator.
L
2.8mm
Fig. 7. Current distribution of the Ant. IV at 36.1 GHz.
20 25 30 35 40
-25
-20
-15
-10
-5
0
5
10
h1
Null 2
Realized Gain (dBi)
Frequency (GHz)
h1 + L2 = 3.28 mm
h1 + L2 = 3.48 mm
h1 + L2 = 3.68 mm
L2
Null 1
20 25 30 35 40
-25
-20
-15
-10
-5
0
5
10
Null 2
Realized Gain (dBi)
Frequency (GHz)
a = 5mm
a = 5.2mm
a = 5.4mm
Null 1
a
(a) (b)
Fig. 8. Realized gain of the proposed Ant. IV against (a) h1 + L2 and (b) a.
20 25 30 35 40
-25
-20
-15
-10
-5
0
5
10
b
a
Null 2
Realized Gain (dBi)
Frequency (GHz)
a = 5.4 mm b = 2.3 mm
a = 5.2 mm b = 2 mm
a = 5 mm b = 1.7 mm
Null 1
20 25 30 35 40
0
20
40
60
80
100
Radiation Efficiency (%)
Frequency (GHz)
Antenna band
Absorptive band
(a) (b)
Fig. 9. (a) Realized gain of the proposed Ant. IV against a & b. (b) Radiation
efficiency of the proposed Ant. IV.
This article has been accepted for publication in IEEE Antennas and Wireless Propagation Letters. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/LAWP.2023.3296150
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
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4
the sake of antenna measurement, the feeding lines are
extended in opposite directions, as shown in Fig. 10(b). The
whole thickness of the proposed antenna array is 1.803 mm,
equal to 0.162 λ0 at 27 GHz. The space of the antenna element is
9.1 mm in both the x-direction and y-direction, equal to 0.82 λ0
at 27 GHz. In order to suppress the surface wave and improve
the realized gain, four sidewalls composed of via holes and
copper lines are added with inner size of 23 mm × 23 mm.
It is worth mentioning that for a filtering antenna design, not
only the in-band but also the out-of-band antenna performance
should be concerned. That means it needs an ultra-wideband
feeding network. To better evaluate both the in-band and
out-of-band performances, two power dividers with different
working band are designed. Fig. 10(d) depicts the S-parameter
of the two designed power dividers. It can be seen that the
return loss of the feeding networks is lower than -15 dB.
Fig. 11 shows the results of the antenna array. The
optimization was performed using high frequency structural
simulator (HFSS), the measurement was accomplished by
Agilent N5227A network analyzer and Satimo system. As seen
in Fig. 11(a), the antenna array has a 22.2% in-band operating
bandwidth. The measured in-band isolation between the two
ports is better than 20 dB. It is worth noting that when one port
is excited, the other port is connected to a 50  load resistor.
With regard to the radiation results shown in Fig. 11(b), the
measured gains are about 11.5 dBi for both the x and y
polarizations, respectively. The average measured efficiency is
about 72% in the antenna band. Also, the measured radiation
efficiencies within the out-of-band are less than 6%. More than
16 dB out-of-band radiation suppression levels are obtained.
The difference between the simulation and measurement results
is mainly due to the fabrication error as well as the variation of
the permittivity and loss of the substrates. In our future work,
we will research on how to improve the fabrication process to
keep the antenna performance unchanged. Fig. 12 shows the
antenna radiation patterns for the two ports. The
cross-polarization levels are lower than -17 dB, while the
backside radiations are less than -16 dB in both E- and H-plane.
Table I tabulates the comparison results of the four
absorptive filtering antennas. Different from the three reported
design using slot, branch and bandstop filters (BSF), the
proposed work used the antenna feeding circuit. It should be
mentioned that the roll-off rates of the antennas are also
compared. Based on the method proposed in [20], the roll-off
rate can be described as the GSI (gain suppression index) which
can be calculated as GSI = GBW3dB/ GBW10dB = (fH3 - fL3) / (fH10
- fL10), where fH3 and fL3 are the higher- and lower-band edge
frequencies, respectively, when the realized gain is 3 dB lower
than the working band gain. Similarly, fH10 and fL10 are the
frequencies when the realized gain is 10 dB lower than the
working band gain. In summary, the millimeter-wave operation,
radiation zero control, dual-polarization function and high GSI
are realized in this work using a simple structure.
V. CONCLUSION
In this letter, a millimeter-wave ME-dipole antenna with
bandpass-type filtering response has been presented. The
design procedure and working principle have been revealed in
detail by circuit analysis. A 2×2 filtering antenna array has been
fabricated and measured for demonstration. It realized a 22.2 %
impedance bandwidth (24.1~30.1 GHz), 11.5 dBi average
in-band gain and more than 16 dB out-of-band radiation
suppression level. The proposed antenna can be easily designed
using the low-cost standard PCB fabrication technique. Above
features make the proposed antenna a good candidate for 5G
millimeter-wave applications.
Sidewall
Antenna elements
Port 1
Port 2
Test connector/Fixture
z
x
y
Mounting hole
Zc=50
Ω
La
λ
g/4
，
Zc=70
Ω
(a) (b)
(c) (d)
Fig. 10. (a) 2 × 2 antenna array structure, (b) feeding structure, (c) fabrication
prototype and (d) simulated results of the power dividers.
20 25 30 35 40
-40
-35
-30
-25
-20
-15
-10
-5
0
5
10
Absorptive band
S-parameters(dB)
Frequency (GHz)
Sim. S11 Sim. S12 Sim. S22
Mea. S11 Mea. S12 Mea. S22
Antenna band
20 25 30 35 40
-25
-20
-15
-10
-5
0
5
10
15
Antenna band
Radiation Efficiency (%)
100
75
50
25
Realized Gain (dBi)
Frequency (GHz)
Port 1 (sim.)
Port 2 (sim.)
Port 1 (mea.)
Port 2 (mea.)
Absorptive
band
0
(a) (b)
Fig. 11. Simulated and measured (a) S-parameters and (b) realized
gains/efficiencies.
-180 -120 -60 0 60 120 180
-40
-35
-30
-25
-20
-15
-10
-5
0
5
10
xoz-plane
Amplitude (dB)
Angle(Degree)
Co-pol (Sim. array) X-pol (Sim. array)
Co-pol (Mea. array) X-pol (Mea. array)
-180 -120 -60 0 60 120 180
-40
-35
-30
-25
-20
-15
-10
-5
0
5
10 Co-pol (Sim. array) X-pol (Sim. array)
Co-pol (Mea. array) X-pol (Mea. array)
yoz-plane
Amplitude (dB)
Angle(Degree)
(a) (b)
Fig. 12. Simulated and measured radiation patterns of the proposed antenna
with Port 1 excited at 27GHz in (a) xoz-plane and (b) yoz-plane.
TABLE I
COMPARISON OF THE FOUR ABSORPTIVE FILTERING ANTENNAS
Design
method
Fre.
(GHz)
Radi. zero
control/No.
GSI
Polarization
[14]
Patch
+slots
+resistor
5-6.5
Yes / 2
~ 0.82
Vertical-
[15]
Quasi-Yagi+
branch
+resistor
2.6-
3.06
No / 2
~ 0.69
Horizontal-
[16]
Slot ant.
+BSF
+resistor
2.85-
3.24
No / 0
~ 0.30
Horizontal-
Pro.
ME dipole
+ ring circuit
24.25-
29.5
Yes / 2
~ 0.76
Horizontal-
/ Vertical-
This article has been accepted for publication in IEEE Antennas and Wireless Propagation Letters. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/LAWP.2023.3296150
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: Xiamen University. Downloaded on July 19,2023 at 01:27:34 UTC from IEEE Xplore. Restrictions apply.

5
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This article has been accepted for publication in IEEE Antennas and Wireless Propagation Letters. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/LAWP.2023.3296150
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