Motion-Modulated Chipless RFID

Abstract

This paper addresses the new type of backscatter communication based on motion-modulated chipless Radio Frequency IDentification (RFID) tags. To clearly explain the concept, the different methods of backscatter communication are classified from a system point of view based on the two fundamental properties of linearity and variation in time. The principle of classical chipless RFID technology, as a non-modulated backscatter method, and the motion-modulated chipless RFID are described with general mathematical demonstrations, while the performance of the two approaches is compared in terms of read range. Motion-modulated chipless RFID is presented as an effective backscatter communication method for identification and sensing of moving objects at large distances. Three different types of motion-induced modulation as phase (Doppler) modulation, polarization modulation, and directional modulation are addressed based on three specially designed moving resonant scatterers. The modulation process in each case is theoretically described, and the performance of the motion-modulated tag is experimentally verified in terms of identification capability and large read range.

Full text

Received 5 October 2022; revised 1 December 2022; accepted 2 December 2022. Date of publication 19 December 2022;
date of current version 6 January 2023.
Digital Object Identifier 10.1109/JMW.2022.3226884
Motion-Modulated Chipless RFID
ASHKAN AZARFAR (Graduate Student Member, IEEE), NICOLAS BARBOT (Member, IEEE),
AND ETIENNE PERRET (Senior Member, IEEE)
(Invited Paper)
University of Grenoble Alpes, Grenoble INP, LCIS, F-26000 Valence, France
CORRESPONDING AUTHOR: Ashkan Azarfar (email: ashkan.azarfar@lcis.grenoble-inp.fr).
This work was supported by the European Research Council (ERC) through the European Union’s Horizon 2020 Research and Innovation Program (ScattererID)
under Grant 772539.
This work did not involve human subjects or animals in its research.
ABSTRACT This paper addresses the new type of backscatter communication based on motion-modulated
chipless Radio Frequency IDentification (RFID) tags. To clearly explain the concept, the different methods
of backscatter communication are classified from a system point of view based on the two fundamental
properties of linearity and variation in time. The principle of classical chipless RFID technology, as a
non-modulated backscatter method, and the motion-modulated chipless RFID are described with general
mathematical demonstrations, while the performance of the two approaches is compared in terms of read
range. Motion-modulated chipless RFID is presented as an effective backscatter communication method for
identification and sensing of moving objects at large distances. Three different types of motion-induced
modulation as phase (Doppler) modulation, polarization modulation, and directional modulation are ad-
dressed based on three specially designed moving resonant scatterers. The modulation process in each case is
theoretically described, and the performance of the motion-modulated tag is experimentally verified in terms
of identification capability and large read range.
INDEX TERMS Backscatter communication, chipless RFID, doppler, motion-modulated chipless RFID,
MTT 70th Anniversary Special Issue.
I. INTRODUCTION
Backscatter communication methods, as it is illustrated in
Fig. 1, can be classified into two main categories as modu-
lated backscatter and non-modulated backscatter techniques
in which the functionality of the transponder can be respec-
tively modeled as a time-variant (TV) and a time-invariant
(TI) system. Then, the both categories are classified based on
linearity, which results in four general classes for backscatter
communication with non-linear time-variant (NLTV), linear
time-variant (LTV), linear time-invariant (LTI), and non-linear
time-invariant (NLTI) transponder. Since this paper is fo-
cused on linear techniques, the non-linear methods, NLTV
and NLTI, are only mentioned by [1] and [2] respectively
to provide a general view, and so they have been shown in
gray color in Fig. 1. For both LTV and LTI transponders,
the magnitude of the reflected electromagnetic (EM) wave
is linearly proportional to the magnitude of the incident EM
wave. However, the main difference between the two linear
techniques (LTV and LTI) is that the backscattered EM wave
from an LTV transponder is modified during the time while
the backscattered wave from an LTI transponder remains in-
variant over time. Basically, an energy source as a modulation
source should be provided in an LTV transponder such that its
properties is modified relative to the data during the time. As
it is shown in Fig. 1, this modulation source is either electrical
or mechanical which can be supplied by the transponder itself
(active LTV transponder) or be captured by the transponder
from external sources available in the environment (passive
LTV transponder). With this proposed general classification in
mind, we explore historically all the linear backscatter com-
munication methods presented up to now, while all of them
can be fitted in this categorization. Moreover, this classifica-
tion is used to accurately address the presented work among
all the other types of backscatter communications.
The earliest backscatter communication system was
invented in 1880 when Alexander Graham Bell realized an
optical backscatter communication link called photo-
phone [3]. In a photophone, the sound waves (human
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256 VOLUME 3, NO. 1, JANUARY 2023

FIGURE 1. Classification of backscatter communication methods.
speech) incident on a flexible mirror caused to vibrate its
reflecting surface and produce a modulation on the reflected
light beam which carries the speech signal. According to
the best knowledge of the authors, the photophone is the
first realized example for LTV backscattering transponder.
With a similar idea but in Radio Frequency (RF) band, a spy
listening LTV transponder called “The Thing” was developed
by Leon Theremin in 1945 [4]. The device was basically a
cavity-loaded antenna in which the impedance seen by the
antenna was modulated with sound waves impinging on the
flexible membrane of the resonant cavity, and consequently,
the backscattered RF waves from the antenna carried the
speech of those close to the device. Afterward, in 1948, Harry
Stockman in his landmark paper [5] classified the different
types of linear backscatter modulation (modulation done
by LTV transponders) as “variable-damping modulation”,
“phase (Doppler) modulation”, “directional modulation”,
and “polarization modulation”. In other words, Stockman
has demonstrated that a “modulated reflector” (or an LTV
transponder) can be realized based on any phenomenon
which changes the scattering properties of the reflector
(scatterer) so that, the backscattered wave is modified in
magnitude, phase, or polarization during the time. Stockman
has explored the concept comprehensively in theory, and he
addressed practical examples for mechanically modulating
LTV transponders realized based on rotating corner reflectors.
It should be mentioned that the mechanical modulation source
in [3] and [4] (vibration) was captured from the sound waves
(passive LTV transponder), while in [5] the rotational motion
was applied to the corner reflectors using electric motors
(active LTV transponder). In addition, the modulation data
in all these cases [3], [4], [5] was directly associated with
mechanical sources, which are speech data carried by the
sound waves in [3] and [4], and the rotational speed in [5].
Although the modulation source in most of the early
backscatter communications was mechanical, the use of elec-
tronic switching elements got widely attracted in afterward
years as an effective method for backscatter modulation with
electrical source. This technique started from 1940 when
the earliest Identification Friend or Foe (IFF) system was
developed using switches which make the dipole scatterers
mounted on the aircraft short or open circuited, and conse-
quently modulate the radar echo of the aircraft like an active
LTV transponder [6]. This was the beginning of the devel-
opment path for chipped RFID technology which today is
used extensively in everyday life and in commercial appli-
cations [7]. The chipped RFID tags basically consist of an
antenna loaded by the electronic switching chip which mod-
ulates the backscattering from the antenna [8]. The electrical
energy required for the modulation in chipped RFID tags can
be supplied by the built-in batteries in semi-passive chipped
RFID tags (active LTV transponder) or can be harvested from
the incident EM waves in passive chipped RFID tags (passive
LTV transponder) [7], [9]. Obviously, the modulation data in
the chipped RFID technology is also associated with (or stored
in) the electronic chip. The great advancement of electronics
in recent decades has paved the way to implement more ef-
ficient and less-cost electronic chips for chipped RFID tags,
however, the fabrication cost for chipped tags is still quite
large compared to the optical barcodes which are used in
massive identification processes. In addition to the cost issue,
chipped RFID tags has a much more complex implementation
compared to barcodes. Furthermore, the large-scale use of
silicon chips has a strong impact in terms of environmen-
tal pollution. As an intermediate solution between chipped
RFID tags and barcodes, chipless RFID technology has been
recently introduced to significantly lower the tag fabrication
cost by removing the electronic chip from the tag structure,
and to relatively alleviate the complexity and environmental
issues [10], [11], [12]. In this technique, the identification
data are included in the physical and geometrical proper-
ties of the chipless tags which are made from passive linear
materials (conductors and dielectrics), and the structural en-
coded data in the tags is translated to their frequency or
time domain backscattering response (frequency-coded tags
based on multiple resonant scatterers and time-coded tags
based on multiple reflectors along a transmission line) [11].
Thus, as there is no time-varying property involved in this
technique, chipless tags can be considered as LTI transponders
which will be discussed in Section II. Although the chipless
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AZARFAR ET AL.: MOTION-MODULATED CHIPLESS RFID
FIGURE 2. (a) Classical communication scenario to identify stationary
frequency-coded chipless tags in a real environment. (b) Interrogating
signal. (c) Tag response. (d) Environment response. (e) Total received
response. (f) PSD of the measured tag response.
approach provides more desirable tags (less expensive, less
complex, and more environmentally compatible) compared to
chipped RFID, the chipless technology faces some challenges
in terms of the coding capacity, reading distance, and reader
complexity [11]. During the last decade, the chipless RFID
has been developed and widely considered by research groups
to introduce efficient coding approaches [13], [14], [15], [16],
robust detection methods [17], [18], [19], [20], and novel
designs for reader architecture [21], [22], [23]. Nevertheless,
compared to the above subjects, fewer studies have focused on
the read range limitation of the chipless tags and the possible
ways to increase the reading distance [24], [25]. In fact, the
achieved read range for chipless tags in most studies is limited
by 1 m which is much lower than what can be reached for
chipped UHF RFID tags.
The study conducted in [26] has demonstrated that the chip-
less read range is fundamentally limited due to the inherent
property of the chipless tags which behave as an LTI transpon-
der. This fact is addressed well in Section II. Indeed, in a real
environment where almost all the objects are stationary and
behave as LTI systems (as same as chipless tags), contribution
of the tag can not be easily separated form the ones due to the
environment, which limits the read range of the chipless tags.
Obviously, the only effective solution for breaking this read
range limitation is to violate the LTI property for chipless tags.
Coming back to the early backscatter modulation tech-
niques, motion (as a time-varying phenomenon) can be used
to modulate the scattering from chipless tags and to realize
LTV transponders which can be detected at larger distances.
As it is illustrated in Fig. 1, this type of LTV transponders
that are called in the following “motion-modulated chipless
tags” have some shared properties between the mechanical
FIGURE 3. (a) Proposed communication process to identify a
motion-modulated chipless tag. (b) Interrogating signal. (c) Tag response.
(d) Environment response. (e) Total received response. (f) PSD of the
differential backscatter power.
LTV transponders and the chipless tags as LTI transponders.
In other words, the identification data for motion-modulated
chipless tags are associated with the structural properties of
the tags (and not linked with the motion properties like what
was presented in [3], [4], [5]), while the motion effect is
just utilized to modulate the backscattered wave such that
the data encoded into the structural properties of the chip-
less tag can be retrieved from the motion-modulated wave.
For this reason, the same chipless RFID tag can belong
to both the “Motion-modulated chipess RFID” and “Linear
Time-invariant” categories. Since Stockman work, only few
papers have investigated motion-modulated chipless tags. One
remarkable example is addressed in [27] where a rotating
chipless tag was read at a distance of 2 m. It should be
mentioned that, for a time-harmonic incident wave, a mov-
ing scatterer (reflector) can modulate the backscattered wave
during the time in terms of different physical parameters
(magnitude, phase, and polarization), while all of them are
translated to an amplitude modulation, a phase modulation, or
a mixed amplitude-phase modulation on the received signal.
This paper reviews different types of motion-modulated
chipless tags which have been recently developed by the
authors. The performance of the motion-modulated chipless
RFID has been demonstrated theoretically and proved ex-
perimentally for all the types. The authors expect that this
approach will open promising prospects for chipless RFID to
identify and to sense moving objects in real scenarios at large
distances.
The paper is organized as follows. Section II presents
the general advantages of the motion modulated chipless
RFID compared to classical chipless RFID in terms of
reader architecture, detection robustness, and read range. The
258 VOLUME 3, NO. 1, JANUARY 2023

phase (Doppler)-modulated chipless tags, the polarization-
modulated chipless tags, and the direction-modulated chipless
tags are respectively demonstrated in Sections III, IV, and V.
Finally, Section VI concludes the paper.
II. CHI PLESS RFID VS. MOTION-MODULATED
CHIPLESS RFID
A. CHIPLESS TECHNOLOGY: LTI TRANSPONDER BASED
ON RCS
1) PRINCIPLE
Fig. 2 presents the backscattering from a frequency-coded
chipless tag and the classical reading process in frequency
domain chipless technology. The example tag consists of three
different-length dipole scatterers (realized on a dielectric sub-
strate) resonating at three different frequencies as f1,f2, and
f3. Most of the time chipless tags consist of resonant scat-
terers and are modeled as simple RLC circuits. Values of R,
L, or C are linked to the structural parameters (dimensions,
spacing, or any other physical parameters) of the tag [11].
However, the most important physical property behind all the
presented models is that the chipless tag can be considered
as an LTI system. In fact, since all the components of the tag
are electromagnetically linear and there is no time-variation
aspect in the tag structure [of course when it is supposed to
be stationary in space, e.g. at r0in Fig. 2(a)], the LTI model
can accurately describe the chipless tag with a very general
view. In order to interrogate the tag, classically, a UWB signal
X(f) in the form of an EM plane wave is sent toward the tag,
which is ideally shown in Fig. 2(b) with a constant-magnitude
spectral representation between fLand fH[21]. It should be
mentioned that, in this example, the chipless tag is configured
such that it preserves the polarization of the incident wave
for the backscattered wave, while it is also possible to have a
depolarizing configuration for the chipless tag [17]. Based on
the LTI model, the frequency domain backscattering response
of the chipless tag can be described as [26]
Y(f)=X(f)·[aH(f)] (1)
where H(f) is related to the classical Radar Cross Sec-
tion (RCS) response [σ(f)] of the chipless tag (from the LTI
model point of view, H(f) is the scattering transfer function
of the tag), and arepresents the round trip propagation loss be-
tween the TX/RX antennas and the tag in free space. With the
assumption of fL<f1,f2,f3<fH, obviously, three peaks
associated with the three resonance frequencies of the dipole
scatterers are observed in the tag response Y(f) [Fig. 2(c)].
The basic frequency coding approach in chipless technology
is to link the presence of each resonant peak in the response
Y(f) to the presence of each corresponding resonant scatterer
on the tag. Classically, this one-to-one association is used to
identify the chipless tag based on its backscattering response.
Although this process is quite straightforward when the tag
is placed in perfectly isolated free space, situation becomes
more complex when the tag is placed in real environment.
In this case, the environment can also reflect and distorts a
fraction of the emitted signal. The reading process is such case
is described in the next part.
2) READING PROCESS BASED ON RCS
The classical chipless reading procedure in real scenarios re-
quires two measurements, one with the tag in the environment
(Exp1) and one without the tag in the environment (Exp2). By
subtracting the result of Exp2 from that of Exp1 [called back-
ground (or empty measurement) subtraction], the response
of the tag can be reached. However, since the background
environment mostly consists of stationary objects made of
linear materials, exactly like the stationary chipless tag, it can
also be modeled with an LTI system, which cause any increase
in power of the transmitted signal will affect both tag and
environment response with the same factor. Assume that in
Exp1, the interrogating signal X(f) is sent to the chipless tag
when it is placed in a real environment with some stationary
objects in the background [Fig. 2(a)]. The total backscattered
signal can be written in frequency domain as [26]
Ytot(f)=X(f)·[aH(f)+E1(f)] (2)
where E1(f) is the transfer function of the environment seen
by the TX/RX antennas and takes into account leakage, cou-
pling and/or reflections over different objects [Fig. 2(e)].
For compensating the environment, the second measurement
(Exp2) is done without the tag and can be described as
Yenv(f)=X(f)·E2(f)(3)
where E2(f) is the transfer function of the environment with-
out the tag [Fig. 2(d)]. Note that E1(f) and E2(f) are generally
not identical due to a modification of the environment between
the two measurements, and/or due to the coupling between
the tag and the antenna (or any other surrounding object).
By subtracting the two measurement results, the measured
response of the tag is obtained as
Ymes(f)=X(f)·[aH(f)] +X(f)·[E1(f)−E2(f)] (4)
where (f)=E1(f)−E2(f) is defined as the transfer func-
tion of the residual environment [26] [Fig. 2(f)]. In practice,
as it is shown in Fig. 2(f), the Power Spectral Density (PSD)
associated to |X(f)[E1(f)−E2(f)]|2is significantly higher
than the reader sensitivity (linked to the noise floor of the
instrument). In addition, since both the tag and the residual
environment are linear systems, increasing the transmitted
energy leads to an increment in the backscattered energy from
both the tag and the residual environment, while the tag-to-
residual environment response ratio remains constant. This
leads to the very significant result that (unlike LTV systems)
as we will see in the next section, it is not possible to increase
the read range by increasing the power emitted by the reader.
3) READ RANGE
The maximal reading distance of a chipless tag can be ex-
pressed as [26]
dc≤4
GtGrλ2σ(f)
(4π)3|(f)|2(5)
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AZARFAR ET AL.: MOTION-MODULATED CHIPLESS RFID
where Gtand Grare respectively the gain of the transmitting
and receiving antennas. Equation (5) clearly shows how chip-
less read range (as an LTI transponder) is related to the RCS of
the tag while it is limited due to the residual environment. To
have an idea, assuming Gt=Gr=8 dB and λ=0.1m,for
a short-circuited half wavelength dipole scatterer with RCS
of σ=−17 dBsm, since the typical value of the residual
environment is |(f)|2=−50 dBm, the maximum read range
is obtained as 63 cm which is independent of transmitted
power. Note that the read range obtained in (5) is totally
different from what is calculated based on the classical radar
equation as
d≤4
P
tGtGrλ2σ(f)
(4π)3Prmin
(6)
in which P
tis the transmitted power and Prmin is the sensitiv-
ity of the receiver. Obviously, contrary to (6), the chipless read
range given in (5) does not depend on the transmitted power
nor the sensitivity of the reader, and it is generally much lower
than the one obtained by (6) [26]. Worth mentioning that, (6)
is widely and accurately used in classical radar applications
where mostly backscattering from the aircraft is not so much
affected by the background environment, as the antennas al-
most see the free space in the sky. This fact is in contrast with
what is usually going on in chipless technology where the tag
should be detected in a crowded background environment.
B. MOTION-MODULATED CHIPLESS: LTV TRANSPONDER
BASED ON DIFFERENTIAL RCS
1) PRINCIPLE
To differentiate the backscattering contribution of the chipless
tag from that of the environment, and so to overcome the read
range limitation of the chipless technology, breaking the time-
invariant property of the chipless tags by utilizing the motion
effect is the main idea in motion-modulated chipless RFID.
Basically, this technique can be applied to identify or sense
properties of any moving object while the modulation type
induced by the movement depends on the motion trajectory
and on how the chipless tag is reconfigured with respect to the
incident wave during the motion. Generally, as it is shown in
Fig. 3(a), since the position and orientation of the scatterers
on the chipless tag are modified during the movement, for a
time-harmonic incident wave, the backscattered wave can be
modulated in terms of magnitude, phase, or polarization over
the time, which results in a mixed amplitude-phase modula-
tion on the received signal.
Fig. 3(a) shows the same frequency-coded chipless tag
as introduced before while it is moving in space along
the trajectory of 
r(t). The tag is assumed in an environ-
ment composed of stationary objects. Similar to the chipless
scenario, to demonstrate the principle of operation, the mov-
ing chipless tag is considered as an LTV system whereas
the environment is still modeled with an LTI system. Since
the moving chipless tag (as an LTV transponder) cannot be
anymore described by transfer functions, the backscattering
process is formulated in time domain, and its frequency do-
main representation is illustrated in Fig. 3(b)–(f). Using the
slow time-varying approximation [28], for a CW input signal
A0cos (2πf0t), the output of an LTV system can be expressed
by A0A(t) cos (2πf0t+φ(t)) where A(t) and φ(t) respec-
tively models the amplitude and phase modulation induced by
the LTV system on the input CW signal. As the time-variation
associated to movements are usually much smaller than the
frequency of input RF carrier, the motion-induced modula-
tion will generate frequency components close around of the
carrier frequency in the spectral representation of the output.
Accordingly, the interrogating signal x(t) in this technique is
considered as a multi-tone CW signal with stepped compo-
nents between fLand fHas
x(t)=
N

i=1
Ai
0cos (2πfi
0t)(7)
with the spectral representation X(f) shown in Fig. 3(b).
Based on the slow time-varying assumption, the response of
the moving chipless tag can be written by
y(t)=
N

i=1
a(fi
0)Ai
0Ai(t) cos [2πfi
0t+φi(t)] (8)
where Ai(t) and φi(t) are respectively the amplitude and phase
modulation induced on each CW component, and a(fi
0)is
the propagation loss at each carrier frequency. Assuming that
three components of X(f) correspond with the three reso-
nances of the chipless tag, the frequency-domain response
of the tag Y(f) can be illustrated as Fig. 3(c) in which the
motion-modulated portion of the tag response is depicted by
green side lobes around the three resonance-associated com-
ponents (red impulses at f1,f2, and f3). Also, the response of
the environment as an LTI system can be written as
yenv(t)=
N

i=1
Ai
0|E(fi
0)|cos [2πfi
0t+∠E(fi
0)] (9)
where E(f) is the transfer function of the environment. Thus,
the spectrum of the environment response Yenv(f) can be
shown as Fig. 3(d), and that of the total received signal Ytot(f)
is obtained by summing Y(f) and Yenv(f), which is presented
in Fig. 3(e). As the main advantage compared to classical
chipless, the motion-modulated part of the tag response will
not be affected anymore by the environment response, which
means the modulated part can be captured directly from the
total received signal Ytot (f) while no background subtraction
is needed.
2) READING PROCESS BASED ON DIFFERENTIAL RCS
The concept of differential RCS (or delta-RCS) has been
originally introduced for chipped UHF tags based on the two
load impedance which are connected to the tag antenna during
the switching process [29]. However, in [30], the definition
of the differential RCS has been generalized for any LTV
transponder based on the spectral analysis of the modulated
backscattered signal. Basically, according to [30], when an
260 VOLUME 3, NO. 1, JANUARY 2023

LTV transponder is excited with a single-tone CW signal, its
differential RCS is directly related to the modulated part of the
backscattered power which is carried by the new frequency
components (around the carrier) generated due to modula-
tion. This modulated power is called differential backscattered
power Pbs d , and for the motion-modulated chipless tag it can
be obtained as
Pbs d (fi
0)=Bi
|Ytot(f)|2df (10)
where band mare the parameters of the carrier-excluded inte-
gration bandwidth Bi=[fi
0−b,fi
0−m]∪[fi
0+m,fi
0+b]
around each carrier fi
0which is shown in Fig. 3(e). The PSD
associated to the differential backscattered power |Yd(f)|2is
shown in Fig. 3(f). The value of Pbs d (fi
0) is non-zero at the
three carriers linked to the resonances of the chipless tag ( fi
0=
f1,f2,f3) and zero for the other ones, which demonstrates the
motion-modulated chipless tag can be identified based on the
differential backscattered power. However, since the received
Pbs d depends on the distance between the antennas and the
tag, it is necessary to define a distance-independent quantity
as differential RCS σdfor general identification goals. Ac-
cordingly, if the distance between the TX/RX antennas and
the moving tag is assumed dm, the differential RCS of the
motion-modulated chipless tag is obtained as
σd(fi
0)=(4π)3(dm)4Pbs d (fi
0)
λ2GrGtP
t
(11)
at each carrier frequency, where P
tis the power of the
transmitted CW components (the same for all components).
Obviously, according to (10) and (11), the profile of σdis
exactly proportional to the differential backscattered power,
and consequently, the motion-modulated chipless tag can be
identified based on the differential RCS. Worth mentioning
that, the ID information retrieved form the differential RCS
is linked to structural properties of the chipless tag (e.g. reso-
nance frequencies here) and not to the motion characteristics,
while the motion effect is just used to modulate the tag ID
on the backscattered wave. Nevertheless, through the reading
process, it is always possible to extract the motion character-
istics (like velocity, acceleration, and period of the movement
for periodic motions) or other properties of the moving object
(like temperature and humidity) as a sensing application.
3) READING ROBUSTNESS AND READ RANGE
In contrast to the classical chipless, as it is shown in Fig. 3(f),
the PSD of the differential backscattered power |Yd(f)|2is
not affected by the residual environment response anymore,
and it is limited just by the noise floor of the reader (reader
sensitivity). In other words, for example, if a CW wave is
sent at f0frequency, for classical chipless technology, the
backscattering from the tag and the environment will be both
received at f0where the environment contribution can blind
the tag response. However, for motion-modulated chipless
tags, the backscattering from the tag is received at the mod-
ulated sidebands around f0[green side lobes in Fig. 3(e)]
while the environment response is still located at f0[pur-
ple impulses in Fig. 3(e)] and it does not affect the tag
response. This fact demonstrates that the reading process for
motion-modulated chipless tags is much more robust against
the environment clutter. However, of course, as the reading
distance increases for motion-modulated tags, the multi-
path effect (multi-reflection of the modulated tag response
from objects in the environment) can degrade the perfor-
mance of the reading process like what is always raised in
telecommunication.
The maximal read range of the motion-modulated chipless
tag is achieved based on the classical radar [(6)], while it
should be modified with the differential RCS (σd) instead of
RCS (σ) as [26]
dm≤4
P
tGtGrλ2σd(f)
(4π)3Prmin
(12)
which clearly shows that the maximum read range of the
motion-modulated chipless tags is not limited and it can be
increased by increasing the transmitted power, using read-
ers with lower sensitivity, and designing motion-modulated
chipless tags with higher differential RCS, although the lat-
ter is mostly dependent on the motion trajectory and its
characteristics.
4) MODULATION SCHEME
The modulation type induced by the motion and also its
modulation index (or modulation depth) [31] (which de-
scribes how much the modulated variable of the carrier signal
(amplitude or phase) varies around its unmodulated level)
depends on how the backscattered wave is modified by the
moving chipless tag in terms of phase, polarization, and mag-
nitude. Nevertheless, specific scatterers (and so chipless tags)
with designed motion trajectories can be proposed to only
modulate the backscattered wave with phase (or Doppler),
polarization, or magnitude, which leads to a pure amplitude or
phase modulation on the received signal. This approach will
be helpful to precisely characterize the tag ID and possibly to
sense the motion properties or the moving object characteris-
tics. In the following sections, we present different types of the
motion-modulated chipless tag which have been employed to
implement different modulation types which can be used for
identification and sensing applications.
III. DOPPLER-MODULATED CHIPLESS TAGS
A. ROTATIONAL MOTION
1) DESCRIPTION
Fig. 4 presents the Doppler-modulated chipless tag based on
the rotational motion [32]. To just modulate the backscat-
tered wave in phase (Doppler effect) during the rotation,
the short-circuited dipole scatterer is rotated such that it is
aligned with the incident wave polarization (aligned along
z-axis with the z-polarized incident plane wave). As it shown
in Fig. 4, in this configuration, the magnitude and polarization
of the backscattered wave will not be modified during the
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AZARFAR ET AL.: MOTION-MODULATED CHIPLESS RFID
FIGURE 4. Phase (Doppler)-modulated chipless tag. The normalized
magnitude of the incident and scattered field is shown as a function of
time during one period of motion (Tm=2π/ωm) respectively in red and
blue color. The phase modulating waveform induced by the motion is also
shown for better interpretation with dashed green line.
rotation, while just its phase is modulated proportional to
the rotational movement. The Doppler modulation induced
by the rotation (as a periodic motion), is observed as a sinu-
soidal time-varying phase (shown by green color in Fig. 4)
βsin (ωmt) where β=4πR/λ is the modulation index in
which Ris the radius of rotation, and ωm=2πfmis the an-
gular frequency of the motion. The sinusoidal phase variation
in time-domain, will be translated to frequency components at
nf
mn=0,±1,±2,...around the carrier f0with the respec-
tive amplitudes of Jn(β), where Jn(x) is the Bessel function
of the first kind of order n. Using the unitary property of the
Bessel harmonics +∞
n=−∞ |Jn(x)|2=1, the analytic expres-
sion of the differential RCS for the rotating dipole scatterer
can be obtained as [32]
σd(f0)=σ(f0)[1 −J2
0(β)] (13)
which is directly related to the RCS of the dipole σ(f0)
with the factor of [1 −J2
0(β)]. Equation (13) shows that the
maximum achievable differential RCS is equal to the RCS
of the dipole, and it can be reached if the rotation radius R
is optimally selected such that, at the resonance frequency,
J0(β)=0.
2) MEASUREMENT
The measurement bench used to verify the rotational Doppler-
modulated chipless tag is shown in Fig. 5(a) composed of
an RF signal generator (HP 8720D) and a spectrum analyzer
(Tektronix RSA3408 A) that are respectively connected to the
closely placed TX and RX antennas (monostatic configura-
tion) [A.H. Systems SAS-571], while both instruments are
synchronized using a 10-MHz reference signal. The trans-
mission frequency ( f0) can be set in-between 1 and 3.5 GHz
with the desired step and the output power (P
t) has been
set to 0 dBm [32]. The chipless tag is implemented by six
different-length dipole scatterers shown in Fig. 5(b) resonat-
ing at 1–3.5 GHz ( fL=1 and fH=3.5 GHz) bandwidth. The
rotation is realized using a motor-driven cylindrical support
with the optimum radius of R=35 mm on which chipless tags
are attached as shown in Fig. 5(c). The measured PSD of the
FIGURE 5. (a) Measurement bench used for identification of the rotating
Doppler-modulated chipless tag. (b) Chipless tag consists of dipole
scatterers. (c) Configuration of the dipoles on the rotating cylindrical
support [32].
FIGURE 6. (a) Measured PSD of the backscattered signal from rotating
Doppler-modulated chipless tag. (b) Measured differential RCS of the
Doppler-modulated chipless tag [32].
backscattered wave from a single rotating dipole is illustrated
in Fig. 6(a) at two different carriers while one of them exactly
correspond to the resonance of the dipole ( f0=fres ) and the
other is at f0=1.4fres. In addition to the clearly observable
rotation induced frequency components at n×22.5 Hz [asso-
ciated with Bessel harmonics Jn(β)] around the carrier in both
cases, the large difference between the amplitude of the mod-
ulated components in two cases (≈20 dB) demonstrates how
the resonance of the dipole can be effectively detected based
on the differential backscattered power, and consequently dif-
ferential RCS, at large distances (1.2 m in this example).
Moreover, note that when the dipole is not attached to the
rotating support, there is no modulated harmonics around the
carrier, which means the σdof the environment will be much
smaller than that of the rotating chipless tag [32].
3) IDENTIFICATION
By sending a sequence of CW carriers between 1 and 3.5 GHz
(fi
0i=1,2,...), the measured differential RCS associated
with each single-dipole tag and also that of a multi-dipole
tag (composed of four dipoles) are calculated based on (10)
and (11) which is shown in Fig. 6(b). Obvious resonance
peaks linked to each dipole scatterer prove the identification
process based on the differential RCS for rotational Doppler-
modulated chipless tags [32].
262 VOLUME 3, NO. 1, JANUARY 2023

FIGURE 7. (a) Read range measurement in a real environment.
(b) Differential backscattered power as a function of distance [32].
4) READ RANGE
The maximal read range of the Doppler-modulated chipless
tag can be estimated using (12) with the calculated differential
RCS shown in Fig. 6(b). To experimentally verify the results
obtained from (12), the maximum achievable read range of
the rotating tag (90 mm length dipole) is measured in a real
environment as it is shown in Fig. 7(a). The measured Pbs d is
plotted as a function of distance in Fig. 7(b) which is in good
agreement with the results theoretically obtained based on
(12). According to the sensitivity of the used reader (spectrum
analyzer), the detection threshold of the tag is measured at
Prmin =−78 dBm which leads to the maximum read range of
almost 10 m with P
t=5 dBm. The achieved maximum read
range for the Doppler-modulated chipless tag outperforms the
classical chipless read range by at least a factor of 10, which
is a great improvement in terms of reading distance as it was
expected [32].
B. VIBRATIONAL MOTION
1) DESCRIPTION
The other Doppler-modulated chipless tag realized based on
vibration is shown in Fig. 4 [33]. The chipless tag is consid-
ered with a rectangular loop scatterer which is aligned such
that the backscattered wave (generated by the fundamental
mode of the loop) preserves the same polarization as the
incident wave (z-aligned). The loop has a one-dimensional vi-
bration along the direction of the incident wave (y-axis) which
induces a sinusoidal phase modulation on the backscattered
wave same as the rotational case, while the modulation index
for the vibration case is defined as β=2πD/λ where Dis the
vibration amplitude. Since the modulation scheme is the same
for the rotating dipole and the vibrating loop, the differential
RCS of the vibrating loop is also expressed by (13) [33]. The
FIGURE 8. (a) Measurement bench used for vibration sensing using
Doppler-modulated chipless tag. Chipless tag composed of rectangular
loop scatterers is shown in the inset. (b) Detected vibration amplitude by
use of Doppler-modulated chipless tag. (c) Averaged detected vibration
amplitude [33].
performance of the vibrational Doppler-modulated chipless
tag in terms of identification and read range is almost similar
to the rotational case [33] which is not presented here to
avoid redundancy. However, it should be mentioned that the β
value in practical vibrations is usually quite small (due to the
small vibration amplitude respect to the wavelength) which
cause to reduce the differential RCS level significantly [see
(13)]. Nevertheless, it is demonstrated in [33] that resonant
scatterers such as a rectangular loop can be utilized to sense
the vibration characteristics (amplitude and frequency) more
efficiently and at larger distances compared to when chipless
tags are not attached to the vibrating surface. Accordingly,
the measurement results related to the vibration sensing using
Doppler-modulated chipless tag is presented in the next part.
2) VIBRATION SENSING
The measurement setup for vibration sensing is shown in
Fig. 8(a) [33]. The vibration is produced using a loud speaker
which is fed by a LF signal generator with the sinusoidal
voltage of V0sin(2πfmt) where fm=68 Hz. Four rectangular
loop scatterers used for the sensing are also shown in the
inset of Fig. 8(a). The loops are attached to the speaker with
a foam support. By measuring the PSD of the backscattered
wave at the resonance frequency of each loop, and using a
least-square estimator based on successive Bessel harmonics
ratios [33], the vibration amplitude is detected at four different
levels of the speaker feed voltage V0=1.25,2.5,3.75,5V.
The corresponding estimated vibration amplitudes are pre-
sented in Fig. 8(b) for the four loops which are compatible
with each other demonstrating the validity of the vibration
sensing performance. Moreover, the average estimated vibra-
tion amplitudes is presented in Fig. 8(c) as a function of V0
which perfectly follows the linear variation of the small signal
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AZARFAR ET AL.: MOTION-MODULATED CHIPLESS RFID
FIGURE 9. Polarization-modulated chipless tag. The normalized
magnitude of the incident and scattered field is shown as a function of
time during one period of motion (Tm=2π/ωm) respectively in red and
blue color. The amplitude modulating waveform induced by the motion is
also shown for better interpretation with dashed green line.
approximation [33]. This example shows that it is possible to
measure vibration amplitudes well below one millimetre with
a very good accuracy while working with a carrier frequency
of a few GHz.
IV. POLARIZATION-MODU LATED CHIPLESS TAGS
A. DESCRIPTION
The polarization-modulated chipless tag is presented in Fig. 9.
The chipless tag is a strip dipole located at xz-plane which
is rotated around y-axis such that it center point coincide
with the origin. The incident plane wave impinges normally
on the rotating strip dipole with a vertical polarization (z-
polarized) [34]. Since the orientation of the tag is varying
during the rotation, the two polarization components of the
backscattered wave will be modulated by the tag. Suppose
that, at the initial moment when the dipole is aligned with
z-axis, the backscattered wave is expressed using the polari-
metric scattering matrix [S]as
Ev
s
Eh
s=Svv Svh
ShvShhEv
i
Eh
i(14)
where vand hpolarization respectively corresponds to zand
xpolarization in Fig. 9. For the described scattering config-
uration, it is shown that the backscattered wave during the
rotation can be written as [34]
[Es]=[(θ)]T·[S]·[(θ)] ·[Ei] (15)
where (θ) is the rotation matrix associated with the time-
varying rotation angle of θ(t)=ωmt. Using (14), the expres-
sion of the two polarization components of the backscattered
wave is obtained as it is indicated in Fig. 9, while the modu-
lation is done on the amplitude of the two field components
with two trigonometric functions of time (marked in green
color in Fig. 9). Note that the phase of the backscattered
wave is not modified by the rotating tag due to the symmetry
of the scatterer. Obviously, for example, if the backscattered
wave is captured by the antenna with the same polarization as
incident wave (vertical polarization), the polarization modu-
lation induced by the chipless tag is observed as an amplitude
modulation on the received signal as it is shown in Fig. 9. The
FIGURE 10. (a) Measured PSD of the backscattered signal from
polarization-modulated chipless tags. Identification is realized based on
the amplitude of the side lobes. (b) Chipless tag formed by strip dipole
scatterers. (c) Configuration of the tags on the rotating circular
support [34].
amplitude modulation in time-domain produce two frequency
components at ±2fmaround the carrier frequency f0in the
spectral domain. Accordingly, the differential RCS of the
polarization-modulated chipless tag can be obtained in terms
of polarimetric scattering parameters as [35]
σd(f0)=4π|Svv (f0)−Shh(f0)|2+4|Svh(f0)|2
8(16)
which is quite similar to (13) as it relates the σdto the static
scattering parameters. However, in contrast to (13) obtained
for Doppler-modulated chipless tags, the differential RCS of
the polarization-modulated tag (16) does not depend on the
motion characteristics (like the rotation radius or vibration
amplitude in case of Doppler-modulated tags) and it is just
determined based on the polarimetric scattering characteristic
of the used scatterer in the chipless tag which is linked to
geometrical properties of the scatterer [34].
B. RESULTS
1) IDENTIFICATION
Using the same measurement bench as Fig. 5(a) (both TX
and RX antennas in vertical polarization), the PSD of the
backscattered wave from polarization-modulated chipless tags
are captured as shown in Fig. 10(a) [34]. Four chipless tags
formed by aluminium-made strip dipoles are used in mea-
surements, which are presented in Fig. 10(b). The transmitted
carrier is set at f0=915 MHz with P
t=0 dBm. Similar
to Fig. 5(a), the tags are attached to a circular support like
what is shown Fig. 10(c), and it is rotated using a motor with
rotational frequency of fm=71 Hz. The tag ID in this case is
associated to the magnitude of the differential backscattered
power (instead of resonance frequencies in spectral behaviour
of the Pbs d for the Doppler-modulated tags). Accordingly, the
amplitude of the two modulation-induced frequency compo-
nents at ±2fm=±142 Hz is investigated to identify the four
designed chipless tags. Obviously, as it is shown in Fig. 10(a),
the highest level of the modulated component is associated
with the biggest tag (Tag 4), and the lower ones are consecu-
tively associated with Tag 3, Tag 2, and Tag 1 [34].
264 VOLUME 3, NO. 1, JANUARY 2023

FIGURE 11. (a) Measured PSD of the backscattered signal from
polarization-modulated chipless tag for different rotational speed. Sensing
is realized based on the frequency position of the side lobes.
(b) Differential backscattered power as a function of distance [34].
2) SENSING
As a sensing application, the rotational speed of the tag
can be also detected based on the modulation-induced fre-
quency components observed in the captured PSD. Note
that as these components appear at ±2fm, the rotational fre-
quency is half of the frequency of the measured components.
Fig. 11(a) illustrates the rotational speed sensing capability
using polarization-modulated chipless tags. As it is shown, the
tag is rotated at different speeds, and the rotational frequency
can be retrieved easily from the location of the modulation-
induced components [34].
3) READ RANGE
Finally, the maximum achievable reading distance of the
polarization-modulated chipless tag is examined experimen-
tally with results presented in Fig. 11(b). The differential
backscattered power from rotating Tag 4 is measured in
an anechoic chamber and in a real environment, and the
measured results are compared with what can be calculated
based on (11). As it can be observed the results are in
good agreement, and the polarization-modulated tag can be
read at distances up to 13 m with P
t=36 dBm and f0=
915 MHz [34].
V. DIRECTION-MODULATED CHI PLESS TAGS
A. PRINCIPLE
The last type of motion-modulated chipless tags is proposed
as direction-modulated tags [36]. Following the two previous
types [phase (Doppler) and polarization modulated tags], this
type of tags should modulate the backscattered wave just
in magnitude (and not in phase or polarization) during the
motion. Although it is not difficult to find moving scatterers
that do not depolarize the reflected wave (like rotating dipoles
in Section III or any other structure which can be perfectly
aligned with the incident field), a moving structure that does
not modulate the backscattered phase (while it has a radial
velocity toward the incident direction) can not be realized
easily. However, since usually the scatterers (when they are
impinged by EM wave) have a specific reradiation pattern
(or RCS pattern) in space which can be strong in one “di-
rection” and weak in the other “direction”, the rotation of
FIGURE 12. (a) Symmetric resonant rectangular loop scatterer proposed
for Doppler suppression. (b) The azimuth distribution of the phase and
normalized magnitude of the scattered electric field for two incidence
angles (normal and oblique) [36].
the scatterer in space can modify the backscattered wave just
in magnitude, provided that the phase (Doppler) modulation
induced by rotation is suppressed somehow. Fig. 12(a) shows
a rectangular loop scatterer centered at the origin and aligned
with z-axis. Assume that the loop is impinged by a z-polarized
wave from two directions: the first one, a general oblique
incidence at φ=φi1and the second one, a normal incidence
at φ=φi2=π/2. Also assume that the carrier frequency is
chosen such that the fundamental resonant mode of the loop
scatterer is excited with the mode current shown in Fig. 12(a).
At the azimuth plane (xy-plane), the reflected wave for all
incidence angles (φ=φi1and φi2) has the same vertical polar-
ization (z-polarization) as the incident wave. In addition, using
a full-wave simulation, the azimuth distribution of the phase
and normalized magnitude of the scattered electric field can
be obtained as shown in Fig. 12(b) for both incidence angles.
The key point is that, for both cases, which means for all the
incidence angles in the azimuth plane, the backscattered field
has the same phase and magnitude distribution. Moreover,
the azimuth phase distribution is perfectly uniform for all
incidence angles. In other words, this specific scatterer can
modify the magnitude of the backscattered wave proportional
to its reradiation pattern when it is rotated around its sym-
metry axis (z-axis), while the phase and polarization of the
backscattered wave are not modified at all. Note that during
the rotation, different points of the loop have a non-zero radial
velocity towards the incidence direction, however, the phase
of the induced current is not modulated due to the resonance
effect, which means the Doppler effect is suppressed by the
resonance of the scatterer [36].
B. DESCRIPTION
Fig. 13 presents the direction-modulated chipless tag which
is designed based on the proposed principle. The resonant
rectangular loop is rotating around its symmetry axis z-axis
and it is impinged by a z-polarized wave. Accordingly, the
magnitude of the backscattered field is modulated propor-
tional to the reradiation pattern of the loop at its fundamental
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AZARFAR ET AL.: MOTION-MODULATED CHIPLESS RFID
FIGURE 13. Direction-modulated chipless tag. The normalized magnitude
of the incident and scattered field is shown as a function of time during
one period of motion (Tm=2π/ωm) respectively in red and blue color. The
amplitude modulating waveform induced by the motion is also shown for
better interpretation with dashed green line.
FIGURE 14. The magnitude and phase variation of the measured
backscattered signal from (a) symmetrically (b) non-symmetrically rotating
loop during one period of rotation. The respective orientation of the loop
at each quarter of the rotation cycle is linked with the waveform by colors
and numbers to illustrate the directional amplitude modulation [36].
mode, whereas the phase and polarization of the reflected
wave are not affected by the rotating loop. According to the
azimuth reradiation pattern of the loop cos(π/2 cos(φ)), the
magnitude of the backscattered field is modulated in time
as it is indicated by the green color function in Fig. 13.
Obviously, this directional modulation is observed as a pure
amplitude modulation on the received signal as it is shown in
Fig. 13 [36].
C. RESULTS
The concept is verified experimentally using the same bench
as Fig. 5(a), while the loop is rotated by the motor with two
configurations as symmetrical and non-symmetrical rotation,
which are respectively shown in Fig. 14(a) and (b). The phase
and amplitude of the received signal in both configurations are
plotted in Fig. 14(a) and (b) as a function of time (during one
period of rotation). As it can be observed, the amplitude of the
received signal in both configurations varies proportionally to
the reradiation pattern of the loop at each rotation cycle. To
clarify that, in Fig. 14(a) and (b), each rotation cycle is divided
into four quarters (depicted in green, red, blue, and black
color) and the link between the orientation of the loop and its
corresponding amplitude signal state is shown by colors and
numbers. This illustration proves the directional amplitude
modulation induced by the rotating loop in both configura-
tions. However, the phase of the received signal is constant
over the time only for the symmetric configuration and it
is sinusoidally varying for the non-symmetric case, which
demonstrates that a pure directional amplitude modulation can
be achieved only when both the symmetry property and the
Doppler suppression due to resonance is utilized simultane-
ously [36].
VI. CONCLUSION AND FUTURE O UTLOOK
The concept of motion-modulated chipless RFID was estab-
lished in this paper. The feasibility of the idea in terms of
read range enhancement was proven theoretically and verified
in the experiment, where the reading distance of more than
several meters was achieved for motion-modulated chipless
tags. Different types of motion-induced modulation were con-
sidered analytically to obtain the expression for associated
differential RCS. Although the concept was addressed for
some familiar periodic motions like rotation and vibration, the
principle holds for any motion trajectory. Accordingly, in the
future, the idea can be applied in more practical cases like
tagged objects carried by a conveyor belt or human hands
holding a tag. Moreover, the presented precise analysis for
modulation schemes and their resultant signal expressions can
be exploited in some sensing and localization scenarios.
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Sci. Rep., to be published.
ASHKAN AZARFAR (Graduate Student Member,
IEEE) received the B.Sc. degree in electrical en-
gineering from Iran University of Science and
Technology, Tehran, Iran in 2014, and the M.Sc.
degree in electrical engineering from the Univer-
sity of Tehran, Tehran, Iran in 2017. He is currently
working toward the Ph.D. degree with the Uni-
versity of Grenoble Alpes, Grenoble INP, France.
His research interests include motion-modulated
chipless transponders, electromagnetic wave scat-
tering, antenna design and wave propagation, and
microwave circuits design.
NICOLAS BARBOT (Member, IEEE) received the
M.Sc. and Ph.D. degrees from the University de
Limoges, France, in 2010 and 2013, respectively.
His Ph.D. degree work with Xlim Laboratory,
Limoges, France, was focused on error-correcting
codes for the optical wireless channel. He also re-
alized a postdoctoral work in joint source-channel
decoding with L2S Laboratory, Gif-sur-Yvette,
France. Since September 2014, he has been an
Assistant Professor with the Université Grenoble
Alpes - Grenoble Institute of Technology, Valence,
France. His scientific background with LCIS Laboratory, Valence, France,
covers wireless communications systems based on backscattering principle
which include classical RFID and chipless RFID. His research interests
include transponders which can not be described by linear time-invariant
systems. This gathers harmonic transponders which are based on the use of
a non-linear component (Schottky diode) or linear time-variant transponders
which are based on the modification of their response in the time domain. He
also places special interests on antenna design and instrumentation based on
these phenomenons.
ETIENNE PERRET (Senior Member, IEEE)
received the Eng.Dipl. degree in electrical
engineering from the Ecole Nationale Supérieure
d’Electronique, d’Electrotechnique, d’Informa-
tique, d’Hydraulique, et des Télécommunications,
Toulouse, France, 2002, and the M.Sc. and Ph.D.
degrees in electrical engineering from the Toulouse
Institute of Technology, Toulouse, France, in 2002
and 2005, respectively. From 2005 to 2006, he
held a postdoctoral position with the Institute
of Fundamental Electronics, Orsay, France. He
was appointed Associate Professor in 2006 and Full Professor in 2022 of
electrical engineering with the University of Grenoble Alpes, Grenoble INP,
France, where he heads the ORSYS Research Group (20 researchers) from
2015 to 2022. From 2014 to 2019, he was a Junior Member with the Institut
Universitaire de France, Paris, France, an institution that distinguishes
professors for their research excellence, as evidenced by their international
recognition. From 2015 to 2020, he was an appointed Member of the French
National Council of Universities. He has authored or co-authored more than
200 technical conferences, letters and journal papers, and books and book
chapters and holds several patents. His works have generated more than 4200
citations. His research interests include wireless communication systems
based on the principle of backscatter modulation or backscattering of EM
waves especially in the field of RFID and chipless RFID for identification
and sensors, electromagnetic modeling of passive devices for millimeter
and submillimeter-wave applications, and advanced computer-aided design
techniques based on the development of an automated codesign synthesis
computational approach. Dr. Perret was a Technical Program Committee
Member of the IEEE International Conference on RFID, the IEEE RFID
TA, and is a member of the IMS Technical Paper Review Committee. He
was the recipient of several awards such as the MIT Technology Review’s
French Innovator’s under 35 in 2013, French Innovative Techniques for
the Environment Award in 2013, SEE/IEEE Leon Brillouin Award for his
outstanding achievement in the identification of an object in an unknown
environment using a chipless label or tag in 2016, IEEE MTT-S 2019
Outstanding Young Engineer Award, Prix Espoir IMT - Académie des
sciences in 2020, and the Grand Prix de l’Electronique Général Ferrié in
2021. He was a Keynote Speaker and the Chairman of several international
symposiums, and also ERC Consolidator Grant in 2017 for his Project
ScattererID.
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