Simple circuits for power electronics

Abstract

New circuits for inverters and power converters are described, that have in common their simplicity.

Full text

Simple circuits for power electronics
Presented at the IEEE seminar of September 2006, Gijón
Translated and updated from the Spanish original (May 2019).
Francesc Casanellas, CEng, MIET, SMIEEE
Abstract: New circuits for inverters and power
converters are described, that have in common their
simplicity.
I. INVERTER [1] [2]
One problem of the high voltage power inverters
is how to drive the MOSFET of the positive side. The
circuit shown in fig.1 is possibly the simplest
configuration, and is extremely fast.
Fig. 1. MOSFET inverter
The inverter has the usual blocking diodes D4, D6
and the anti-parallel diodes D5, D8.
When Q3 is turned on, Q2 gate is shorted to ground
through R4, which limits de discharge current and
dampens oscillations. C3 is charged to +12 V through
D2. The negative dV/dt in C4 creates a current that is
diverted by D3 and blocks Q1.
When the gate in Q3 goes low, Q3 turns off. The
current C4·dV/dt flows through the base of Q1 and this
transistor charges the drain capacitance or Q3 and the
gate capacitance of Q2, that turns on. R3 allows that Q1
stays on, after the transient, compensating the leakage
current of Q3.
For a short while (some tenths of nanoseconds),
specially when Q3 turns off and Q2 turns on, both
MOSFETS are conducting. A small inductor L1 limits
the current. Often a simple ferrite bead can be used.
D1, R1 and C2 remove the energy stored in L1.
Component values correspond to a three-phased
inverter for 0,75 kW, 230 V motors. The same
configuration has been used at 800 V D.C. and
thousands of invereters have been built. More details
in ref. [1]. A variant with an additional transistor for a
higher gate current is described in [2].
II. SIMPLE FLY-BACK CONVERTER WITH
LOW E.M.I. AND VERY FAST RESPONSE
Fig. 2 shows a very simple fly-back converter, but
with very good performance. It works in the boundary
of the discontinuous mode, so the output diodes start to
conduct when its current is null, and there is no
problem caused by the recovery charge. Turn on of the
main transistor Q1 is relatively slow, so the EMI are so
low that usually no filters are necessary.
Another advantage of this circuit is the main
transistor Vcb0 of 1200V, so it is much rugged than most
power supply integrated circuits.
It has extremely fast response and inherent stability.
Feedback gain can be adjusted via R5.
Fig. 2. SMPS Fly-back.
Base current of Q1 starts through R1-R3-R6. The
voltage drop through LED1 (a red LED that acts as a
low voltage zener and at the same time shows that the
supply is working) is higher than Vbe of Q1 plus Vf of
D6. The base current of Q1has a positive feedback
thanks to the auxiliary winding of the transformer and
D4. Collector current increases (I = V·t/L) until the
voltage in the shunt resistor R13 triggers the transistor
pair Q2-Q3, connected as a thyristor. The emitter
voltage is the voltage drop in R13 plus the voltage in
D4, so Q1 turns off very fast. Then the voltage in the
auxiliary winding inverts and through LED1, the
current in R1-R3 is diverted from the base of Q1. When
all the energy in T1 is transferred to the secondary, this
voltage stops and the cycle restarts.
The output voltage is controlled by Z1, OC1 and
R11. The feedback gain is controlled by the value of
R5.
The components shown in the schematics are
chosen for a 10 W supply. But with the selected power
transistor (2 A), much higher powers can be achieved.

For better output voltage accuracy, a TL431 can be
used, instead of Z1.
IV. SIMPLE MOSFET FLY-BACK
CONVERTER
Figure 3 shows a fly-back converter similar to the
previous one, but using a MOSFET. The advantage is
that the frequency can be higher and the transformer
smaller. The operation is quite similar to the previous
one, but the LED and diodes are replaced by C6, which
avoids that the auxiliary winding short-circuits the
starting current from R3.
Fig. 3. Fly-back with MOSFET
V. “QUASI-RESONANT FLY-BACK
CONVERTER USES A SIMPLE CMOS IC [3].
Fig. 4 shows a fly-back power supply which
has very low noise and uses a simple CMOS 4093
integrated circuit for the control. The electrical noise of
a converter is produced mainly when current is
switched on: diode recovery and charge of parasitic
capacitances create high di/dt which is the cause of
noise. This converter achieves very low noise level
switching current on very slowly at nearly zero voltage.
The converter works in the boundary between
discontinuous and continuous mode, and switches on
when the drain voltage is at its lowest value.
To avoid to work with low gate voltages, which
would cause excessive MOSFET losses, ZD1 conducts
and enables the input gate of the 4093 when the voltage
is high enough. When the supply starts, the auxiliary
non isolated winding through D3 keeps the gate input
high.
When the MOSFET is on, current increases linearly
until the base of Q5 starts to conduct and this transistor
turns the MOSFET off. Then the flyback operation
starts, and the primary energy charges the output
capacitors. During this stage, D5-R6 keep Q5
conducting and the MOSFET off. When the energy has
discharged, D5 stops conduction as well as the
secondary diodes (so no recovery problems exist).
The constant time of C5-R5 keeps the MOSFET off
for a while: The output capacitance of the MOSFET
(plus parasitic capacitance of the primary) resonates
with the primary inductance and the voltage decreases.
C5-R5 are chosen to allow the MOSFET to turn on
when the voltage has reached the minimum value. The
values shown are valid only for a particular case.
In this way not only turn on losses are reduced, but
also the electrical noise. Voltage regulation is done in
a traditional way using a TL431. Optocoupler current
is added to the shunt current.
Because the MOSFET turns on when current is 0,
the gate resistor may be very high, so parasitic

capacitances are charged slowly, further reducing
switching noise.
The circuit around Q4 is optional and can be used
in most power supplies. Its kills the current glitch when
Q3 turns on, being much effective than the usual RC,
and allowing a very low duty cycle at low loads.
VI. MOSFETS IN SERIES FOR HIGH
VOLTAGE
Using MOSFETs in series is easier than it may
seem. Fig. 5 shows a 250 W forward converter working
up to 400 V input. It uses two 500 V MOSFETs. As the
silicon area is proportional to the square of the voltage,
it is more effective to use two 500 V MOSFETs than
one of 800 V.
Fig. 5. Converter with MOSFETs in series.
Q1 supports the supply voltage, and Q2 the
reflected voltage that appears when both MOSFETs
turn off.
When Q1 is on, R1 keeps C1 charged and Q2
conducting. When Q1 turns off, its voltage drain raises
fast and when ZD1 conducts Q2 gate is fully
discharged and Q2 turns off too. The drain voltage of
Q1 is limited to Vdc by ZD1 and D1.
When Q1 turns on, C1 charges Q2 gate. The circuit
can be used with a fly-back converter.
This topology is very usefull for power supplies
working at 380-415 V A.C. input (up to 600 V D.C.)
Fig. 6 shows an 80 W buck converter working at
1700 V input. It uses two 1000 V MOSFETs in series.
When both MOSFETs are off, Q2 gate is kept to Vdc/2
by R1-R8 and R9-R10 avoid any voltage imbalance
due to the drain-source leakage current.
The snubbers allow the share the voltage between
the two MOSFETS during commutation times. C1 has
the same function as in the previous circuit.
Fig. 6. 1700 V converter
VII. GLITCH SUPPRESSOR [4]
Diode reverse charge and parasitic capacitances
cause a peak of current when the main switch turns on.
Fig. 7. Glitch suppressor.
The typical solution is an RC filter as shown in fig.
7A. The RC widens the glitch, as well as decreasing its
height. On light loads, the pulse is so narrow that its
height is lower than that of the filtered glitch, so the
current stops prematurely and the converter becomes
unstable.
The circuit shown in fig. 7B shorts the initial glitch
allowing a better stability at low loads and better
current limit when the output voltage changes.
Fig. 8. RC and active glitch suppressors

Fig. 8 shows the current signal in R2, for high and
low load of a converter working in discontinuous
mode, and the input signal to the IC with an RC and an
active glitch suppressor.
Component values that work for most converters
are C1 = 47 pF, R3 = 4,7 kΩ, R4 = 1 kΩ.
VIII. SLOPE COMPENSATION
The classic recommended circuit is shown in fig.
9A. But it has some drawbacks: the base current of Q2
interferes with the oscillator circuit and it can only be
used with controllers having the oscillator ramp
accessible.
The circuit shown in fig. 9B is universal and does
not charge the oscillator. The ramp is created by R2-
C6. When the MOSFET turns off, the ramp is reset
through D1. For a full explanation of the slope
compensation and the calculation of the components,
see ref. [5].
Fig. 9. Slope compensation
IX. OVERVOLTAGE PROTECTED LINEAR
POWER SUPPLY [6]
Fig. 10 shows a power supply that was designed for
cars, where the formidable pulse of "load dump"
(battery disconnected at high speed) requires high
energy voltage suppressors. In spite of trying to absorb
the energy of the overvoltage, this circuit shuts off
when the overvoltage appears. It withstands without
problem 200 V pulses: when an overvoltage appears,
ZD2 conducts, and turns off Q1, through Q2.
Without any additional component for
compensation, the regulator is very stable and has a
very fast response.
The output voltage can be adjusted a little by
changing the value of R5. R1 and C3 limit the slope of
the input pulse, and are not indispensable.
Fig. 10. Overvoltage protected power supply
The quiescent current is lower than 1 mA. If a lower
current is required, R5 value can be increased. For
example, with 10 kΩ, the quiescent current is about 120
µA. But then the output voltage drops to about 3,7 V
and the zener voltage of ZD1 has to be changed to 5,6
V. An advantage of this power supply over other ones
is that the output can sink current through ZD1 and Q3.
So protection diodes to the positive can be used.
X. POWER SUPPLY FOR LEDS.
Most circuits to supply power LEDs from a
relatively high voltage, have the configuration shown
in fig. 11, where U1 works at constant frequency and
turns off Q1 when the current reaches a certain limit.
Fig. 11. Current supply for LEDs
This circuit cannot allow to control the LED
current with high accuracy. Most integrated circuits
(UC38…, UCC38…, HV9910, etc.) have 10%
accuracy of the current limit. The normal accuracy of
the inductor is 10% and when the ripple changes, as the

current its limited by its peak, the average current
changes.
To increase the current accuracy, we have inverted
the usual configuration. In fig. 12 the LEDs are in the
negative side, and thanks to the way D1 is connected,
all the LED current goes through the shunt resistor
R1. The average current can be obtained with a
simple filter. Most PWM controllers have a reference
voltage with 2% accuracy. Even a simple comparator
with hysteresis can be used as a control, as shown if
fig. 13.
Fig 12. Accurate current controlled LED supply.
XI. FREQUENCY MODULATION ("JITTER")
TO DECREASE E.M.I.
A trick to decrease the interference level in power
converters is to use a modulation frequency to spread
the interference energy through a frequency band
In most power supplies fed from A.C. mains this
can be made very simply with only two components,
as shown in fig. 14. R1 and C1 modulate the
frequency at the 100 Hz of the ripple supply voltage.
The ripple, increases with the load, so the
modulation increases with the load, which is
favourable.
Fig. 14. Modulation of the switching frequency.
REFERENCES
[1] F. Casanellas. “Circuit makes simple high voltage
inverter”. EDN, May 27, 2004.
[2] Patents: US4802075, EP0274336
[3] F. Casanellas. “Quasi resonant converter uses a simple
CMOS IC”. EDN, April 15, 2004.
[4] F. Casanellas. “Deglitcher for more stable switching
power supplies”. Electronics World, November 1996.
[5] F. Casanellas. “New slope compensation method
stabilizes switchers”. EDN, March 25, 2016
[6] F. Casanellas, “Power supply meets automotive-
transient voltage-specs”. EDN, September 18, 2008.
Fig. 13. Accurate current controlled LED supply using a comparator.

MORE PAPERS ON POWER ELECTRONICS:
“Compensación del Tiempo Muerto y de los Retardos en Inversores y Amplificadores de Potencia Clase D (Dead
time compensation)”. Eurofach Electronica, núm. 329, 2004, pp. 60-63.
https://www.researchgate.net/publication/262562172
“Losses in PWM inverters using IGBTs”. IEE Proceedings -Electr. Power Appl., Vol. 141, No. 5, September
1994.
“Improvements in phase shift full bridge converters” https://www.researchgate.net/publication/281242618
“Synchronous rectification for forward converters”
https://www.researchgate.net/publication/265602097
https://www.academia.edu/28085536
“Cálculo de los elementos pasivos de conmutación en un inversor con SCR”. Mundo Electrónico, 1984, no. 144.
Update and English translation:
“Calculation of the passive components and the commutating current in an assisted turn off inverter”.
Seminario anual de automática, electrónica industrial y automación, Gijón, 2006.
https://www.researchgate.net/publication/262562262
https://www.academia.edu/28085534