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JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 16, NO. 11, NOVEMBER 1998 1953
A Femtosecond Code-Division Multiple-Access
Communication System Test Bed
H. P. Sardesai, C.-C. Chang, and A. M. Weiner
Abstract—This paper reports comprehensive experimental re-
sults on a femtosecond code-division multiple-access (CDMA)
communication system test bed operating over optical fiber in
the 1.5
m communication band. Our test bed integrates to-
gether several novel subsystems, including low-loss fiber-pigtailed
pulse shapers for encoding–decoding, use of dispersion equalizing
fibers in dispersion compensated links for femtosecond pulse
transmission and also in femtosecond chirped pulse amplification
(CPA) erbium doped fiber amplifiers (EDFA’s), and high-contrast
nonlinear fiber-optic thresholders. The individual subsystems
are described, and single-user system level experimental results
demonstrating the ability to transmit spectrally encoded fem-
tosecond pulses over a 2.5-km dispersion compensated fiber link
followed by decoding and high contrast nonlinear thresholding
are presented.
Index Terms— CDMA, dispersion compensation, nonlinear
thresholding, pulse-shaping, ultrafast phenomena.
I. INTRODUCTION
T
O meet the demand for high-speed and high-capacity
communications, multiple-access schemes are necessary
which allow multiple users to access the network simultane-
ously by sharing the same fiber-optic transmission medium.
For long distance communication, wavelength division multi-
plexing (WDM) and time-division multiplexing (TDM) have
been extensively explored and have shown impressive perfor-
mance results [1]–[6]. For local area network (LAN) appli-
cations, optical code-division multiple-access (CDMA) com-
munications systems [7]–[20] are also being investigated, in
addition to the more traditional WDM and TDM schemes.
Each data bit in an optical CDMA system is coded with a
code that is unique to a particular user, and multiple users
are accommodated by assigning different minimally interfering
codes to different user pairs. Several different minimally
correlated code-sequences exist in traditional digital communi-
cation systems that can be implemented in the optical domain
making optical CDMA suitable for multi-user operation, the
main constraints being the fidelity of the encoding–decoding
operation for a user pair and the consequent successful de-
Manuscript received November 20, 1997; revised July 10, 1998. This work
was supported by the National Science Foundation under Grants ECS-96
26967 and ECS-9312256.
H. P. Sardesai was with the School of Electrical and Computer Engineering,
Purdue University, West Lafayette, IN 47907 USA. He is now with Ciena
Corporation, Linthicum, MD 21090 USA.
C.-C. Chang was with the School of Electrical and Computer Engineering,
Purdue University, West Lafayette, IN 47907 USA. He is now with Bell
Laboratories, Lucent Technologies, Holmdel, NJ 07733 USA.
A. M. Weiner is with the School of Electrical and Computer Engineering,
Purdue University, West Lafayette, IN 47907 USA.
Publisher Item Identifier S 0733-8724(98)08307-8.
tection in the presence of interference from other users.
CDMA is well suited for bursty network environments, and
optical CDMA has the advantage of using optical processing
to perform certain network applications like addressing and
routing. The asynchronous nature of data transmission can
simplify network management and control. Hence, due to the
advantages of optical processing, asynchronous transmission,
and the capability of multiple-access in a bursty environment,
optical CDMA appears to be an interesting possibility for LAN
applications. On the other hand, the technologies required
for implementing optical CDMA systems are significantly
less mature and may be inherently more complex than those
required for TDM or especially WDM systems.
Several different optical CDMA schemes have been pro-
posed [7]–[20], based on different choices of sources, coding
schemes and detection. Two reviews of optical CDMA are
given in [14] and [15]. Optical CDMA schemes may be
classified according to the choice of coherent versus in-
coherent processing, coherent (modelocked pulses) versus
incoherent (e.g., amplified spontaneous emission) broadband
optical source, and encoding method (time-domain versus
frequency-domain, amplitude versus phase). Schemes based
on incoherent processing (summing of optical powers) and
broadband incoherent (noise) sources are generally the easiest
to implement. However, as in radio spread spectrum, coherent
processing based on manipulation of optical fields, which can
be made to sum to zero, is needed for good suppression of
multiple-access interference. Note that coherent processing is
possible even for systems using incoherent noise sources, e.g.,
coherence multiplexing approaches based on interferometric
techniques; however, recent analyzes have shown that optical
beat noise becomes a major factor limiting the capacity of such
systems [14]. Here, we experimentally investigate an ultrashort
pulse optical CDMA scheme based on spectral phase encoding
and decoding of coherent modelocked pulses [7], [15]. Note
that for ultrashort pulse CDMA, multiple-access crosstalk and
optical beat noise are essentially synonymous. A theoretical
analysis of the cross-talk limited performance of this approach
indicates the potential for CDMA systems with capacities from
tens to perhaps (100 Gb/s, depending on how short a pulse
width and how long a code length can be maintained during
system operation [7], [15].
A block diagram of the ultrashort pulse CDMA scheme
configured for LAN applications is shown in Fig. 1. In the
transmitter, femtosecond laser pulses are spectrally encoded by
a pseudorandom phase code that transforms the femtosecond
pulses into picosecond duration pseudonoise signals. Each user
0733–8724/98$10.00 1998 IEEE
1954 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 16, NO. 11, NOVEMBER 1998
Fig. 1. Block diagram of the femtosecond CDMA test bed.
(transmitter) is assigned a unique phase code by which it
encodes all its data bits, and this phase code is chosen to be
minimally interfering compared to the phase codes assigned
to every other user in the system. Different users can be
connected in a simple broadcast and select type architecture
where all transmitters are connected to all receivers by a
passive star coupler. Each receiver thus receives encoded
data bits transmitted by every transmitter in the network,
but the decoder in any one receiver matches the phase code
of only one transmitter. Hence, only the encoded data bits
of one transmitter that are intended for a particular receiver
get properly decoded back to a femtosecond pulse, and the
encoded data bits transmitted by all other transmitters remain
as improperly decoded pseudonoise signals. A nonlinear fiber-
optic thresholder then performs the task of distinguishing
between the correctly decoded femtosecond signal and the
incorrectly decoded picosecond interference. In this CDMA
scheme, each transmitter may operate at moderate data rates
(e.g., on the order of 1 Gb/s), but with multiple-access higher
overall data transmission rates may be achieved.
In this paper we demonstrate for the first time to our
knowledge that all the operations required for femtosecond
pulse CDMA, namely spectral encoding, fiber transmission
with dispersion compensation, spectral decoding, and non-
linear optical thresholding, can be accomplished with good
fidelity in an integrated system. Some of our preliminary
results were described earlier [20]. In the course of con-
structing an ultrashort pulse CDMA “test bed,” we have
developed several ultrashort pulse CDMA component and
subsystem technologies that may also have broader application
to ultrafast optical communications and have begun to assess
component technology limitations that may impact overall
system performance. Note that in our optical CDMA test bed
we currently demonstrate single-user operation consisting of
one transmitter and one receiver. We currently encode every
laser pulse, although a real system would use a modulator that
can modulate the femtosecond laser pulses according to the
incoming data stream (e.g., using on-off keying). The encoded
pulses are propagated over a 2.5-km fiber link which uses
dispersion compensating fiber (DCF) to compensate both the
second and most of the third-order dispersion of the standard
telecom fiber. Dispersion compensation is crucial since the
encoding–decoding operation requires linear and substantially
dispersion free pulse propagation. The transmission distance
in femtosecond optical CDMA is limited mainly by the effec-
tiveness of the dispersion management scheme used. Although
we currently demonstrate propagation over only 2.5 km, by
using programmable third-order dispersion correction in the
encoder we have achieved almost dispersion free transmission
for sub 500 fs pulses. This opens the possibility of having
longer propagation distances of tens of kilometers. In this
paper we give a comprehensive description of our femtosecond
CDMA experiments, both at the subsystem and system level,
including new results on encoding–decoding of femtosecond
pulses after propagation through fiber, programmable dis-
persion compensation of coded-pulses, system results after
higher order dispersion compensation, and system results for
encoding–decoding with codes of different lengths.
In the following, Section II describes the different individual
component technologies used to construct the CDMA test
bed and presents subsystem level results. Section III presents
system level results with all the CDMA subsystems connected
together. In Section IV, we discuss some limitations of our
ultrashort pulse CDMA implementation as revealed by the
experiments. In Section V we conclude.
II. CDMA C
OMPONENT TECHNOLOGIES
The CDMA link integrates together several novel subsys-
tems including femtosecond lasers and amplifiers, femtosecond
fiber pig-tailed pulse shapers for encoding and decoding,
femtosecond dispersion compensation and ultrafast nonlinear
thresholders. This section describes the various building blocks
and presents subsystem level experimental results.
A. Femtosecond Lasers and Amplifiers
Ultrashort optical CDMA requires femtosecond laser pulses
as they provide the wide bandwidth and phase coherence
necessary for the encoding–decoding operation. The lower
limit on the shortest femtosecond pulse that can be used is
placed by the effectiveness of the dispersion compensation
scheme over the transmission distance. The upper limit on
the longest pulse that can be used is placed by the minimum
bandwidth required to code the ultrashort pulses, and the short
pulsewidth required for effective high contrast thresholding.
Due to these conflicting requirements for optimal operation of
the different subsystems that make up the CDMA system, a
pulsewidth of a few hundred femtoseconds was chosen for
our experiments. Although several different techniques for
femtosecond pulse generation in the 1.55
m communication
band exist, a passively mode-locked fiber laser was used due
to its advantages of ease of construction and compatibility
with all-fiber systems. Our femtosecond laser source is a
passively mode-locked stretched-pulse all-fiber ring laser [21]
that generates
62 fs pulses with a bandwidth of 60 nm. The
laser output is externally filtered by a bandpass filter resulting
in
275 fs pulses, with an average power of 40 W, at a
repetition rate of
30 MHz. A complete description of the
laser construction may be found in [33]; only the intensity
autocorrelation traces and spectra from the laser before and
after the bandpass filter are shown in Fig. 2 demonstrating
clean transform-limited laser operation. Note that although
our 30 MHz pulse source was sufficient for characterizing the
fidelity of the different CDMA operations, higher repetition
rate sources would be required in a practical system.
Two amplifiers are used in the ultrashort pulse optical
CDMA link. First, a preamplifier directly after the filtered laser
SARDESAI et al.: FEMTOSECOND CDMA COMMUNICATION SYSTEM TEST BED 1955
(a) (b)
(c) (d)
Fig. 2. Autocorrelation data and power spectra at the output of the femtosec-
ond laser before and after bandpass filtering. (a) spectrum before filtering, (b)
autocorrelation (
62 fs intensity FWHM) before filtering, (c) spectrum after
filtering, and (d) autocorrelation (
275 fs intensity FWHM) after filtering.
source to compensate for the insertion loss of the encoder and
the link, and second a postamplifier after the decoder to ensure
adequate power for nonlinear thresholder operation. Although
the two amplifiers were designed to provide different levels
of amplification and output saturation powers, their general
construction is quite similar. Fig. 3 shows a schematic of
the preamplifier. Chirped pulse amplification technique [22],
[23] is used to reduce the nonlinear effects in the amplifier.
Input pulses from the laser are first stretched by passing them
through
60 m of single mode fiber. They are then amplified
by about
15 dB by passing through 18 m of erbium
fiber which serves as the gain medium. The erbium fiber is
pumped by 980 nm light from a Ti-Sapphire laser coupled to
the amplifier through a WDM coupler. Monitors for both the
pump and input signal are provided to measure input signal
and pump power levels. Isolators at the input and output are
used to reduce any feedback effects that can reduce the gain
and output power. The preamplifier was designed to generate
1.2 mW of power at 1550 nm when pumped by 24 mW
of pump power. The amplified pulse is compressed by using
a dispersion-compensating fiber that compresses the pulse to
its transform limited value. The amplifier produces
375 fs
pulses, the pulse being broadened due to gain narrowing effects
in the erbium fiber. The output pulses are taken from the
90% port of a 10–90% output coupler. The postamplifier
follows the same construction as the preamplifier, with the
longer length (
25 m) of erbium fiber to give higher output
powers. The postamplifier is designed to deliver up to
20
mW of output power when pumped by
130 mW of pump
power. At higher output power levels, in addition to gain
narrowing in the amplifier, we see some nonlinear effects in
Fig. 3. Chirped pulse amplification erbium doped fiber amplifier.
Fig. 4. Fiber-pigtailed programmable liquid crystal modulator pulse shaper.
the compressed pulses and the pulses are further broadened to
between
600–900 fs.
B. Femtosecond Fiber Pig-Tailed Pulse
Shapers for Encoding and Decoding
The femtosecond optical CDMA scheme is based on en-
coding and subsequent decoding of ultrashort light pulses. We
accomplish this encoding–decoding operation by using fem-
tosecond pulse-shapers [24]–[27] which offer high-resolution
pulse shaping, programmability and the flexibility to apply
arbitrary phase codes of different code lengths. Encoding and
decoding of femtosecond pulses was previously demonstrated
at a visible wavelength [24], [25], but in that arrangement,
the encoding–decoding operation was performed by using
two fixed conjugate phase masks placed successively in the
same pulse-shaper. We demonstrate here programmable en-
coding–decoding operation at 1.55
m communications wave-
length in two separate pulse-shapers, and also demonstrate
encoding–decoding when the two pulse-shapers are separated
by a 2.5-km fiber link. We have also fiber-pigtailed our pulse
shapers, which increases the ease with which we can either
connect the pulse-shapers in the whole system, or disconnect
them for individual measurements. Fiber-pigtailing also has the
advantage that the pulse-shapers have to be aligned only once
during the initial construction phase. We have achieved a low
fiber-to-fiber insertion loss of only 5.3 dB. To our knowledge,
these are the first experiments demonstrating femtosecond
pulse-shaping operation at 1.55 microns using fiber-pigtailed
pulse-shapers, and also the first demonstration of femtosecond
encoding–decoding operation using such pulse-shapers.
The experimental arrangement of the pulse-shaper is shown
in Fig. 4. In the pulse-shaper, collimated light from the input
fiber pigtail is first diffracted off a grating (1100 lines/mm)
and the different spectral components are then collected and
focused by an achromatic lens (focal length
190 mm). The
incident angle and diffraction angles are approximately 43 and
75
, respectively. At the focal plane of the lens, the spectral
components of the input pulse are linearly spatially separated.
1956 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 16, NO. 11, NOVEMBER 1998
(a) (b)
(c) (d)
(e) (f)
Fig. 5. Cross-correlation data and power spectra at the output of a single
pulse shaper. (a) Output pulse (
440 fs) with constant phase applied to
the pulse shaper, (b) output pulse with 31 element
-sequence coding, (c)
output pulse with 63 element
-sequence coding, and (d)–(f) power spectra
corresponding to (a)–(c), respectively.
The liquid crystal modulator (LCM; CRI model SLM-128)
is used to set the spectral phases to a length 31or length 63
-sequence (MS) pseudorandom phase-code (which encodes
the pulses into 10–20 ps wide pseudonoise bursts) or held
constant leading to essentially unchanged uncoded pulses. The
LCM has a fully programmable linear array of 128 pixels
with 100
m center-to-center pixel spacing, and individual
pixels can be controlled by applying up to 4096 different drive
levels resulting in phase shifts from 0–
[26]. A length 31
MS, for example, consists of a pattern of “1”’s and “0”’s
31 bits in length that is accommodated by the 128 pixels of
the LCM by assigning four pixels to each bit. For MS bits
equal to one, the phases of the corresponding LCM pixels are
set to
radians; for bits equal to zero, the phase is set to
0 radians. The rest of the pulse-shaper consists of a second
matched achromatic lens and grating which reassembles the
different spectral components into a single collimated output
beam which is then coupled back into an output fiber pigtail.
Note that two half waveplates were used, before and after the
LCM, respectively, since the polarization state for operation
of the LCM was orthogonal to that required by the gratings
for optimum diffraction efficiency.
Fig. 5 shows measurements of time-domain output wave-
forms and spectra where the LCM is programmed for either a
constant phase (no encoding) or length 31 or 63
-sequence
phase codes. The time-domain measurements are intensity
cross-correlation data using unshaped pulses from the laser as
a reference; to a good approximation they represent the actual
output temporal intensity profiles. The holes in the spectrum
seen in Fig. 5(e) and (f) are related to diffraction effects arising
from the frequency components of the input pulse which fall
at 0–
transitions of the LCM’s in the pulse shaper [22], [27].
Each individual frequency component of the input pulse has
a finite spatial extent at the mask plane (as determined by the
input beam diameter), which may cause the different spatial
regions of one particular frequency component to see different
phase retardations. The output fiber pigtail acts as a spatial
filter that samples the frequency dependent diffraction pattern
from the LCM. This results in phase to amplitude conversion
that leads to the observed dips in the spectrum. These effects
are more pronounced for the length 63 MS due to a larger
number of 0–
transitions.
We can model these diffraction effects using a simple theo-
retical analysis published previously [27], [28]. The response
of the pulse shaper can be characterized in the frequency
domain by
(1)
where
and are the Fourier transforms of the
input and output electric fields, respectively, and
is the
complex frequency response of the linear filter acting on the
femtosecond pulses.
can be related to the actual physical
masking pattern with complex transmittance
(i.e., the
spatial phase pattern on the LCM) by
(2)
Here
is the spatial dispersion of the pulse shaper with units
cm (rad/s)
1
and is the radius of the focused electric field
beam profile at the masking plane (for any single frequency
component). Expressions for
and in terms of the pulse
shaper parameters and input beam profile are given in [27].
Assuming that the Gaussian mode selected by the output fiber
is matched to the mode from the input fiber, as is the case in
our experiments, (2) completely accounts for diffraction effects
arising in the pulse shaping process. Equation (2) shows that
the effective filter in the frequency domain is the mask function
convolved with the intensity profile of the beam. The
main effect of this convolution is to limit the full-width at half-
maximum (FWHM) spectral resolution
of the pulse shaper
to
Physical features on the mask smaller
than
are smeared out by the convolution, and this limits
the finest features which can be transferred onto the filtered
spectrum. One consequence of this picture is that wavelength
components impinging on mask features which vary too fast
for the available spectral resolution are in part diffracted out
of the main beam and hence not coupled into the output fiber.
This leads to the phase-to-amplitude conversion evident in
Fig. 5. Improved spectral resolution can be achieved, e.g., by
increasing the input beam size to decrease
Fig. 6 shows a comparison of the experimental power
spectrum for coding with a length 31 MS with a simulation
based on (1) and (2). The simulation parameters (
and )
were found by matching the locations and widths of dips in
experimental and simulated power spectrum for a very simple
setting of the LCM where pixels 20, 64, and 110 were set for
phase shift, with all other pixels set for zero phase shift.
This resulted in
cm (rad/s)
1
(0.75 mm/nm)
and
m. Using these same parameters, excellent
SARDESAI et al.: FEMTOSECOND CDMA COMMUNICATION SYSTEM TEST BED 1957
(a)
(b)
Fig. 6. Experimental (a) and simulated (b) power spectra for coding with a
length-31
-sequence spectral phase code using a single phase shaper.
agreement between the actual and simulated spectra for length-
31 MS coding is obtained, as seen in Fig. 6. Similar agreement
is obtained for coding with
-sequences of other lengths
and for encoding–decoding experiments (discussed next), in
all cases using the same values for
and .
Encoding and decoding experiments are performed by tak-
ing pulses exiting from the first pulse-shaper (encoder) and
inputting them through a fiber pigtail into an identically
constructed second pulse-shaper (decoder). The encoder and
decoder are matched to within a pixel accuracy of the LCM’s,
as measured by a 0.08 nm resolution optical spectrum analyzer.
Note that due to aberrations in the pulse shaper arising due to
the very large diffraction angles, as well as the interactions
of polarization mode dispersion effects in the fiber compo-
nents with polarization sensitive devices (e.g., gratings) in
the pulse shaper, the overall output pulse after two pulse
shapers is broadened to
500 fs. Fig. 7 shows experimental
cross-correlation data and corresponding output spectra for the
encoding–decoding operation for length 31 MS phase codes.
Fig. 7(a) shows normalized intensity cross-correlation data for
an uncoded pulse where a constant phase is applied to the
LCM’s in both the encoder and the decoder. Fig. 7(b) shows
cross-correlation data for a properly decoded pulse (PDP) for
31 element MS encoding–decoding when the phase codes of
the two LCM’s match, and Fig. 7(c) shows an improperly
decoded pulse (IPDP) when the phase codes on the two LCM’s
do not match. Note that the vertical axes in Fig. 7(b) and 7(c)
are normalized to the peak intensity of Fig. 7(a). Fig. 7(d)–(f)
show the output spectra corresponding to Fig. 7(a)–(c) respec-
tively. It can be seen from Fig. 7(a) and (b) that although
the encoding–decoding process restores the pulse-width of
the PDP to its original uncoded value, its peak intensity is
reduced to
60% of that in Fig. 7(a). As before, diffraction
effects inherent in the pulse-shaping process cause a decrease
in the peak intensity and the appearance of a pedestal in the
decoded pulse. They are also responsible for the holes seen in
the spectra of Fig. 7(e) and (f). Note that the spectrum of the
IPDP shows more holes. This is because the MS in the encoder
and decoder do not match resulting in a larger number of 0–
(a) (b)
(c) (d)
(e) (f)
Fig. 7. Cross-correlation data and power spectra after encoding–decoding
with two pulse shapers for 31 element
-sequence coding. (a) Output pulse
(
500 fs) with constant phase applied to both pulse shapers, (b) properly
decoded output pulse (
510 fs), (c) improperly decoded output pulse, and
(d)–(f) power spectra corresponding to (a)–(c), respectively.
transitions. Note also that the holes observed in the spectrum
of the PDP [Fig. 7(e)] are wider than those observed in the
spectrum of the encoded pulse [see Fig. 5(e)]. This occurs due
to phase to amplitude conversion at the same pixel position
(and therefore at the same wavelength) arising independently
in both the encoder and decoder. Similar trends were observed
for length 15, 63, and 127
-sequences. In all cases, the
decoded pulse exhibited a main peak duration comparable to
that of the uncoded case and intensity substantially above a
lower intensity pedestal. However, the holes in the spectrum,
the drop in peak intensity, and the overall energy loss in the
decoding process became more severe for longer code lengths.
Table I shows the peak intensity and the energy of the PDP,
normalized to the case of constant spectral phase in the encoder
and decoder, for length 15, 31, and 63 codes. The experimental
results are in excellent agreement with simulations, using the
same values for
and as previously. The comparison
shown in Table I is perhaps the most demanding test of
(2) to date, in terms of the complexity of the experimental
waveforms. The excellent results suggest that this simulation
procedure can be used to predict coding-decoding performance
for a broad range of experimental parameters. Based on the
data in Table I, we have selected length 31 and 63 MS codes
for our system studies, although longer codes (e.g., 127) would
be desirable if sufficient spectral resolution were available.
C. Femtosecond Dispersion Compensation
Transmission of femtosecond pulses over kilometer
distances requires the simultaneous compensation of both
the quadratic dispersion and most of cubic dispersion of
the input pulse. In femtosecond optical CDMA dispersion
1958 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 16, NO. 11, NOVEMBER 1998
TABLE I
N
ORMALIZED PEAK POWER AND ENERGY OF PROPERLY DECODED PULSES FOR
CODING WITH LENGTH 15, 31, AND 63
-SEQUENCES.THE POWER AND ENERGY
ARE NORMALIZED TO THE
POWER AND ENERGY OF PULSES PASSING THROUGH
AN
ENCODER AND DECODER PAIR EACH SET FOR CONSTANT
SPECTRAL PHASE
Normalized
peak power
Normalized energy
-sequence
length
Experiment Theory Experiment Theory
15 73.2% 74.1% 83.1% 78.8%
31 59.0% 65.5% 74.6% 75.0%
63 28.5% 35.7% 46.4% 48.8%
compensation is necessary for two reasons. First, since the
CDMA scheme needs linear pulse transmission due to the
phase sensitive encoding–decoding operation, we cannot
use soliton propagation. Second, uncompensated dispersion
will severely degrade the contrast between properly and
improperly decoded pulses. Several dispersion compensation
schemes applicable to femtosecond pulse transmission have
been demonstrated before that can compensate the chromatic
dispersion of standard single-mode fibers [29]–[33]. Our dis-
persion compensation scheme based on the use of dispersion
compensating fiber (DCF) [34] to compensate the quadratic
dispersion and most of the cubic dispersion of standard single
mode fiber (SMF), has been detailed previously [31]–[33].
Such an SMF-DCF fiber link has much lower third-order
dispersion than conventional dispersion shifted fiber. Further,
by applying a cubic phase to the pixels of the programmable
liquid crystal (LCM) in the encoder, we can almost completely
remove the small residual third-order dispersion of the SMF-
DCF link resulting in essentially distortionless transmission
of sub-500 fs pulses over 2.5 km of optical fiber [33].
To our knowledge, these were the first experiments of
dispersion compensation on a femtosecond time scale using
dispersion compensating fiber [31] and the first demonstration
of almost dispersion free transmission by applying residual
phase correction via a programmable pulse-shaper [33]. In
addition to its applicability in femtosecond CDMA systems,
this dispersion compensation scheme can be used in any other
transmission scheme that uses ultrashort pulses.
Fig. 8 shows intensity cross-correlation data for pulses at
the input and output of the 2.5 km SMF-DCF link. The link
is composed of
2060 m of SMF and 445 m of DCF fiber,
dispersion optimized by adjusting the lengths of the individual
fibers to give the shortest output pulse. Fig. 8(a) shows the
input pulse and Fig. 8(b) shows data for the same pulse after
the 2.5 km link. The output pulse is broadened to
580
fs with some small oscillation in the tail indicating residual
positive third-order dispersion. In contrast, we estimate that
the pulse after propagating down the 2 km length of SMF only
would broaden to
200 ps. The residual dispersion is further
compensated by applying an appropriate phase variation across
the pixels of the LCM in the encoder, resulting in almost
complete dispersion compensation as seen in Fig. 8(c). The
phase pattern applied to the LCM [see Fig. 8 (d)] is discretely
sampled over the entire 128 LCM pixels, but since the phase
difference between the first and last pixel is quite small
(
2.1 ), the sampling can be considered almost continuous.
This leads to the almost exact phase correction of the residual
(a) (b)
(c) (d)
Fig. 8. Cross-correlation data for femtosecond dispersion compensation. (a)
Input pulse to the 2.5 km link, (b) output pulse from the link when a constant
phase is applied to the LCM, (c) output pulse with cubic phase correction
applied to the LCM, and (d) the cubic phase correction function applied to
the LCM pixels.
third-order dispersion in the link and thus to almost complete
restoration of the output pulse.
The pulse shaper can also be programmed for simultaneous
dispersion compensation and decoding (or encoding). This is
accomplished by summing (modulo
) the phases needed for
decoding and for dispersion compensation. Fig. 9 shows inten-
sity autocorrelation data for properly and improperly decoded
pulses using a length 63 MS code for the case of (a) only a few
meters of fiber between encoder and decoder, and (b), (c) 2.5
km dispersion compensated link connecting encoder–decoder,
either (b) without or (c) with one of the LCM’s also used to
trim out the residual phase from the fiber link. Note that in
each case the amplitude of the PDP is normalized to unity, and
a small coherence spike is observed at the origin for the IPDP
as is expected in autocorrelation traces for pseudonoise bursts
[35]. Compared to Fig. 9(a), we can see from Fig. 9(b) that
residual dispersion in the fiber link has broadened the main
peak and reduced the contrast between the PDP and IPDP
autocorrelations. In this case the fiber link was adjusted so
that small amounts of residual quadratic and cubic dispersion
were both present. By programming an LCM for simultaneous
decoding and dispersion compensation as in Fig. 9(c), the
duration of the autocorrelation peak and contrast ratio between
the PDP and IPDP is restored to that observed with only
a few meters of fiber. The ability to perform decoding and
programmable fine tuning of the dispersion compensation in
the same module relaxes to some degree the precision with
which the fixed dispersion compensator must be set.
D. Ultrafast Nonlinear Thresholders
Optical CDMA receivers need a thresholding device to dis-
tinguish between properly decoded femtosecond pulses and the
SARDESAI et al.: FEMTOSECOND CDMA COMMUNICATION SYSTEM TEST BED 1959
(a)
(b)
(c)
Fig. 9. Intensity autocorrelation data for spectral encoding and decoding
separated by fiber, using length-63
-sequences. Both properly (dashed line)
and improperly (solid line) decoded pulses are shown. (a) Only fiber pigtails (a
few meters) separate encoder and decoder. (b) 2.5 km dispersion-compensated
links, without phase trimming by LCM. (c) 2.5 km dispersion-compensated
link, with phase trimming by LCM.
Fig. 10. Schematic of the nonlinear thresholder.
equally energetic improperly decoded picosecond interference
signals. This required discrimination is achieved by exploiting
nonlinear frequency shift effects in optical fibers. We use two
nonlinear effects in optical fibers, namely, nonlinear self phase
modulation [36]–[38] and nonlinear Raman effects manifested
as the soliton self frequency shift [39], [40]. In both of these
effects when a high intensity femtosecond pulse is propagated
in an optical fiber, the output pulse exhibits frequency shifts
(away from its mean input frequency), the exact nature of
the shift depending on the particular nonlinear process. The
lower peak power longer duration interference signals do not
exhibit any significant changes to their frequency spectrum.
Fig. 10 shows the block diagram of the nonlinear thresholder,
which is combination of a suitable length of optical fiber
followed by a long wavelength pass spectral filter. The long
wavelength pass filter is one half of a pulse shaper, with
the LCM replaced by a knife-edge mounted on a translation
stage. This arrangement allows us to change the filter cutoff
wavelength by simply moving the spatial position of the
(a) (b)
(c) (d)
Fig. 11. Power spectra at the output of the nonlinear thresholder for two
different thresholder fibers. (a) Output power spectrum for
nm DSF thresholder fiber for coded pulses, (b) output power spectrum for
nm DSF thresholder fiber for uncoded pulses, (c) output power
spectrum for
nm DSF thresholder fiber for coded pulses, and (d)
output power spectrum for
nm DSF thresholder fiber for uncoded
pulses. For (a) and (b) the average power is 0.44 mW and for (c) and (d) the
average power is 1.84 mW.
knife-edge. The spectrally filtered pulse exiting the filter is
focused into a photodetector. The combination of spectral filter
and photodetector converts any frequency shifts occurring in
the thresholder fiber into amplitude variations which can be
detected by the photodetector.
We earlier demonstrated two different thresholder designs
for a stand-alone thresholder by propagating coded and un-
coded pulse through the nonlinear fiber and obtained high
contrast thresholding after the output filter [41], [42]. In the
integrated system described later, such coded and uncoded
pulses would correspond to improperly decoded and properly
decoded pulses respectively. In the first design [41], both
the uncoded femtosecond pulses and the coded interference
signals were propagated through a dispersion shifted fiber
(DSF) whose zero dispersion wavelength coincided with the
center wavelength of the transmitter laser (
1559 nm). Non-
linear self-phase modulation effects cause the spectrum of
the femtosecond signal pulse to split and spread on either
side of the zero dispersion point while the low intensity
picosecond interference signal remains at its original spec-
tral position. Fig. 11(a) and (b) show power spectra at the
output of the thresholder fiber for coded and uncoded pulses
respectively clearly revealing the differences between the two.
The pulses were coded using a length 63
-sequence. The
long wavelength pass filter at the output transmits the shifted
portion of the uncoded pulse and rejects the unshifted coded
pulse. High contrast ratios of 30 dB were obtained using
500 m of fiber at average power levels of 0.44 mW for
1569 nm cutoff wavelength of the output spectral filter (see
Fig. 12). Fig. 12 also shows the variation of the contrast ratio
for the various cyclic shifts of the 63 element
-sequence
(thus representing different codes or interfering users) for two
1960 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 16, NO. 11, NOVEMBER 1998
Fig. 12. Variations of the contrast ratio at the output of the nonlinear
thresholder for two different positions of the long wavelength pass filter for
coding using the cyclic shifts of a length 63
-sequence. The thresholder
fiber has
nm and the average power in the fiber is 0.44 mW.
different cutoff wavelengths of the output spectral filter. The
contrast ratio is seen to be only minimally affected by the
particular choice of the
-sequence. The energy conversion
efficiency (ratio of energy detected after the spectral filter to
energy in thresholder fiber) for this thresholder was about
10%. The main advantage of this design is the lower average
power required for obtaining high contrast ratios. The main
disadvantage is the requirement to closely match the zero
dispersion wavelength of the thresholder fiber to the center
wavelength of the source laser.
In the second design [42], a DSF fiber with zero dispersion
wavelength less than the center wavelength of the source laser
(
nm) was used so that the optical spectrum lies
entirely in the anomalous dispersion regime of the fiber. The
nonlinear Raman effect and the resulting soliton-self-frequency
shift cause the mean wavelength of the high intensity properly
decoded signal to shift to longer wavelengths, while the low
intensity interference signal remains at its original spectral
position. Fig. 11(c) and (d) show power spectra at the out-
put of the thresholder fiber for coded and uncoded pulses
respectively for this design. Again, a properly positioned long
wavelength pass filter can transmit the shifted signal and reject
the interference signal giving high contrast thresholding. A
nonlinear thresholder having 36 dB contrast ratio for 1.84
mW average power in the thresholder fiber was demonstrated
using 340 m of this dispersion shifted fiber for 1577 nm cutoff
wavelength of the output spectral filter. The main advantage
of this design is the flexibility of choosing the zero-dispersion
wavelength, although it is achieved at the expense of higher
average powers required for high contrast thresholding. To
our knowledge, these were the first experiments demonstrating
nonlinear thresholding operation in optical fibers with high
contrast ratios.
III. S
YSTEM RESULTS
We have so far discussed the various CDMA subsystems
and presented experimental data for their performance. We
(a)
(b)
Fig. 13. Power spectra after the thresholder fiber for 31-element
-sequence encoding–decoding and propagation over a 2.5 km link.
(a) Properly decoded pulse and (b) improperly decoded pulse. The
thresholder fiber is the
nm DSF and the average power is
2.4 mW.
now discuss system level results for a single transmitter-single
receiver experiment including encoding, fiber propagation,
decoding, and thresholding. For single user operation, the
following three parameters will chiefly determine the system
performance. First, the fidelity of the encoding–decoding op-
eration with the 2.5-km fiber link in-between the encoder and
decoder. Second, the effectiveness of the dispersion compen-
sation scheme for coded pulse propagation with and without
residual third-order dispersion correction, and finally, the con-
trast ratio after the nonlinear thresholder between a properly
and improperly decoded pulse.
Fig. 13 shows power spectral data at the output of the
thresholder for PDP and IPDP for 31 element
-sequence
(MS) phase coding. In the system experiments, the thresholder
fiber with zero dispersion wavelength at 1559 nm was used,
primarily because it requires lower average powers to give
high contrast thresholding. By comparing Fig. 13(a) and (b)
we note that the spectrum of the PDP has split to either side
of the zero dispersion point. The peak at 1530 nm observed on
both the spectra is due to the amplified spontaneous emission
from the erbium doped fiber amplifier in the receiver. Fig. 14
shows the encoding–decoding autocorrelation data after the
decoder and the corresponding power spectral data after the
thresholder for 63 element MS encoding–decoding, clearly
demonstrating CDMA operation for longer code lengths. The
contrast ratios (defined as the ratio of the energy of the PDP
to that of the IPDP) after spectral filtering in the nonlinear
thresholder are plotted Fig. 15 for length 31 and length 63
MS phase coding. The cutoff wavelength of the spectral
filter is
1573 nm. The horizontal axis in the figure is the
pump power applied to the EDFA in the receiver, and the
corresponding variation of the average signal power in the
thresholder fiber would be from
1 to 2.5 mW. As seen in the
SARDESAI et al.: FEMTOSECOND CDMA COMMUNICATION SYSTEM TEST BED 1961
(a) (b)
(c) (d)
Fig. 14. Autocorrelation data after the decoder and power spectra after the
thresholder for 63 element
-sequence encoding–decoding. (a) Autocorre-
lation of properly decoded pulse, (b) autocorrelation of improperly decoded
pulse, (c)–(d) power spectra after the thresholder corresponding to (a)–(b),
respectively. The average power in the thresholder fiber is
2.3 mW.
figure, a slight increase in the contrast ratio is observed when
the LCM provides third-order dispersion correction. Note that
the contrast ratio for 31 element MS encoding–decoding is
larger than that for 63 element MS encoding–decoding, and the
difference between the two is more prominent at lower pump
powers. This is because the PDP has a larger peak power (and
also larger average power) for 31 element MS coding than for
63 element MS coding (see Table I). At lower pump powers
the EDFA gain is fairly constant, resulting in the amplified
properly decoded pulse for 31-element coding having a larger
peak power, and therefore higher frequency shifts, and higher
contrast ratios. At higher pump powers, gain saturation effects
come into play and the differences in the contrast ratios are
smaller. The contrast ratio in both cases is limited by the long
wavelength ASE components of the EDFA in the receiver.
Compared to the earlier thresholding experiments using only a
single pulse shaper for encoding (and only a single amplifer),
ASE is a more serious issue in the integrated system, since
two EDFA’s are employed. Note also that for 31 element MS
coding, the contrast ratio curve is quite flat over the entire
range of average powers in the thresholder fiber indicating
that the contrast ratio is not very sensitive to the exact value
of the average power. This is important as it gives some design
margin for constructing the receiver amplifier especially under
multiple user operation.
To get around the ASE limitation, we have two choices.
First, we can install a bandpass filter after the receiver EDFA
that eliminates the long wavelength ASE components. Second,
we can engineer the spectral filter in the thresholder and set
its cutoff wavelength to a much longer wavelength (
1573
nm) effectively blocking out as much long wavelength ASE
as possible. Note that in the second approach we also reduce
some of the signal from the properly decoded pulse. This is
Fig. 15. Contrast ratios at the output of the CDMA test bed for 31 element
and 63 element
-sequence encoding–decoding. Solid lines correspond to
no cubic phase correction in the encoder LCM and dotted lines correspond
to cubic phase correction in the encoder LCM. The cutoff wavelength of the
spectral filter in the nonlinear thresholder is 1573 nm in each case.
Fig. 16. Contrast ratios at the output of the CDMA test bed for 31 element
and 63 element
-sequence encoding–decoding for (
1573 nm) cutoff
wavelength of the nonlinear thresholder (both shown by solid lines). The
dashed-dot line is for a 31-element
-sequence user when the interfering user
has a 63-element
-sequence phase code. The dotted line is for a 63-element
-sequence user when the interfering user has a 31-element -sequence
phase code.
however not a serious limitation as long as we have sufficient
signal for detection, and the reduction of the long wavelength
ASE is greater than the reduction in the properly decoded
signal. Using the second approach we increased the contrast
ratio of the CDMA test bed to 27.5 and 25 dB for 31 element
and 63 element
-sequence coding respectively (compared
with
18 and 15 dB in Fig. 15) as shown by the solid lines
in Fig. 16. Note again that the horizontal axis in the figure
is the pump power applied to the EDFA in the receiver, and
the corresponding variation of the average signal power in the
thresholder fiber would be from
1.5 to 2.75 mW.
We also tested one more variation of the code length
dependence of the contrast ratio, namely when the interfering
1962 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 16, NO. 11, NOVEMBER 1998
user has encoded its data using a different length code-
sequence than the intended user. We can now have some
users who encode their data with length 31 MS transmitting
information over the same optical channel with other users
who encode their data with length 63 MS. Note that in optical
CDMA the intended receiver has to be provided with a-
priori information about the exact nature of the code sequence
of the transmitter. Hence having different users encoding
their data bits with different length MS does not add any
more complexity to the system. When the CDMA receiver
decodes an incoming interference signal, its output remains
as a low intensity pseudonoise burst irrespective of the exact
nature of the code length of the interfering user. This can
be observed from Fig. 16, where it should be noted that the
contrast ratio has actually increased for length 31 MS coding
when the interfering user has its bits coded with a length 63
MS. This can be attributed to a combination of two factors.
First, an interfering user with 63 element MS coding has a
longer temporal spread of its encoded pulse [also compare
Fig. 5(b) and (c)] than an interfering user with 31 element
MS coding. Hence after decoding, a length 31 MS-length 63
MS improper decoding results in a longer duration improperly
decoded pulse than a length 31 MS-length 31 MS improper
decoding. Since the thresholder is a nonlinear device, a longer
duration improperly decoded pseudonoise signal has relatively
less spectral shifts than a shorter duration improperly decoded
pseudonoise signal. (Note that the absolute spectral shifts in
either case are much less than that for a properly decoded
pulse). This explains the increase in contrast ratio. The second
factor for this increase is the slightly higher loss in the decoder
pulse shaper for 63 element MS coding than for 31 element
MS coding. This would cause the decoder output to have
different average powers depending on the specifics of the
encoding–decoding process. The contribution due to this effect
is expected to be small as measurements at the EDFA output
for the two cases (i.e., length 31 MS-length 31 MS and length
31 MS-length 63 MS improper decoding) have shown only
5% difference in average powers. The contrast ratio for 63
element MS encoding–decoding is likewise higher than that
for 63 element coding-31 element decoding. This also appears
to be related to the temporal and intensity characteristics of
the improperly decoded pulses. Note that for a given MS,
translating the bit pattern by one-bit results in a new MS that is
orthogonal to every other MS obtained by such bit translations.
Hence, a length 63 MS can accommodate 63 possible users. A
combination of length 63 and length 31 MS thus increases the
number of addresses that can be assigned to users, and also
shows the robustness of the optimal CDMA system when the
interfering users have different types of codes.
IV. D
ISCUSSION
Although the component technologies and the single-user
system results in principle show potential for true multi-user
operation, several factors may affect the practical implementa-
tion of the CDMA system in a multi-user environment. Some
of the issues which would affect practical femtosecond CDMA
operation are discussed in the following, with reference where
applicable to our experimental results.
1) The current single user experiments running at pulse
rates of
30 MHz required on the order of 1 mW for
high contrast operation of the fiber nonlinear thresh-
older. In multi-user networking each receiver will see a
sample of each of the multiple-access signals; therefore,
the required postamplifier saturation power scales with
number of users (as well as bit rate). For 30 users with
ON–OFF keying at 1 Gb/s per user, each postamplifier
will need to amplify to
500 mW. Although this is
possible, for most applications such an amplifier will
be too costly for use on a per node basis. Therefore,
thresholding devices that can operate at lower power lev-
els than in the current experiments are required. It may
be possible to achieve some power reduction by using
longer thresholder fibers or longer pulsewidths. Other
technologies based on nonlinearities in guided wave
optoelectronic devices may also offer some potential for
lower operating powers [43], [44].
2) We have demonstrated distortionless transmission of
sub-500 fs pulses over a 2.5 km dispersion compensated
fiber link. Although programmable dispersion compen-
sation in the encoder or decoder allows fine tuning of
the overall dispersion balance, nevertheless each fiber
link in the CDMA system will still need rather precise
setting of its large fixed dispersion compensator. Greater
precision will be needed if shorter pulses or longer fibers
are desired, and this may ultimately limit the usable
pulse width or fiber span.
3) Assuming adequate power budgets are available, the
overall capacity in this ultrashort pulse CDMA scheme
scales inversely with pulse width (for fixed code length)
and increases strongly with increased code length (for
fixed pulse width) [7]. Here we have demonstrated
operation with 500-fs pulses and code lengths of 31
and 63. However, to obtain capacities in the range of
tens of Gb/s to
100 Gb/s or above, one needs code
lengths in the range from 127 to 511 and shorter pulse
widths (
100–300 fs). For the current pulse width longer
code lengths (at least up to 127) should be possible by
improving the pulse shaper spectral resolution. Shorter
pulse widths would allow a greater increase in code
length, since more spectrum is available for coding.
To maintain shorter pulse widths, gain narrowing in
the amplifiers, which is a significant limitation in the
current experiments, must be avoided. Additionally, sub-
stantially longer code lengths would require LCM’s (or
other modulator array technologies) with more than the
current 128 pixels.
4) The accumulated nonlinearity in the transmission fibers
must remain small in order to avoid degrading the
decoded pulses. On the other hand, if the transmit-
ted power is too low, the power requirements of the
nonlinear thresholder will place additional demands on
the receiver amplifier. We have performed simulations
and experiments showing that uncoded pulses can be
transmitted with average powers up to the
1mW
SARDESAI et al.: FEMTOSECOND CDMA COMMUNICATION SYSTEM TEST BED 1963
level in the current setup (at the same pulse repetition
rate) before nonlinearities become evident [45]. Coded
pulses (as here) have lower peak intensities and therefore
are less susceptible to nonlinearities. Based on these
considerations, at present we do not expect nonlinearities
in the transmission channel to be a serious limitation.
However, further study is needed to fully assess non-
linearity limits for femtosecond pulse transmission in
dispersion compensated links with large pulse stretching
and compression ratios.
5) In a
user system, the broadcast star architecture will
lead to a factor of
splitting loss not present in the
current single user experiments. This loss would have
to be offset either by using a more powerful source
(compared to the 40
W after spectral filtering currently)
or through additional amplification.
6) Even though we have demonstrated relatively low-loss
filter pigtailed operation, the encoding–decoding device
in its present form requires the use of bulk gratings
and lenses which limits its use in practical applications.
In the future this part of the system will have to be
miniaturized, perhaps taking advantage of integrated
wavelength division multiplexing technologies. Exper-
iments demonstrating simple pulse shaping operation
have been reported both using integrated acoustooptic
tunable filters [46] and arrayed waveguide gratings [47],
and a miniaturized and packaged pulse shaping setup
used for gain equaliztion of amplified WDM systems
was demonstrated in [48].
V. C
ONCLUSION
We have presented a detailed description of a femtosec-
ond optical CDMA scheme. On the subsystem level, three
main component technologies, namely, femtosecond encod-
ing–decoding, femtosecond dispersion compensation, and ul-
trafast nonlinear thresholding, have been developed and char-
acterized. The high fidelity femtosecond encoding–decoding
obtained for length 63 and length 31
-sequences has shown
the potential for true multi-user operation. Femtosecond dis-
persion compensation, especially with residual third-order
dispersion correction, should extend the propagation distance
to over 10 km. The high-contrast thresholding should al-
low good discrimination against multi-access interference,
although lower operating power would be desirable. On a
system level, the ability to propagate a coded pulse and decode
it with a 27.5 dB contrast against interference has the potential
to extend optical CDMA beyond the single-user operation
demonstrated here. Similarly, the high contrast ratios obtained
for different length
-sequence coding makes it possible to
add more addresses than would be available with a fixed
single-length
-sequences. In conclusion, we have demon-
strated for the first time the ability to propagate femtosecond
optical pulses from CDMA transmitters to receivers in an
integrated system including all the required operations needed
for femtosecond pulse CDMA. In the future we plan to use this
system as a test bed to investigate CDMA system performance
during multiuser operation.
A
CKNOWLEDGMENT
The authors would like to gratefully acknowledge A.
Vengsarkar of Lucent Technologies, NJ, for providing the
dispersion compensating fiber, V. DaSilva and M. Newhouse
of Corning Inc., Corning, NY, for providing dispersion shifted
and erbium doped fibers, and I. Duling of Naval Research
Laboratory, Washington, DC, for helpful discussions related to
erbium amplifiers. They would also like to thank A. Emmanuel
and S. Shen for calibrating the LCM’s, and D. Leaird for
technical assistance.
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