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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 8, AUGUST 2007 1771
Design of Vertically Stacked
Waveguide Filters in LTCC
Tze-Min Shen, Chi-Feng Chen, Ting-Yi Huang, and Ruey-Beei Wu, Senior Member, IEEE
Abstract—This paper proposes four-pole quasi-elliptic function
bandpass waveguide filters using multilayer low-temperature
co-fired ceramic technology. The vertical metal walls of the
waveguide resonators are realized by closely spaced metallic vias.
Adjacent cavities are coupled by a narrow slot at the edge of
the common broad wall or an inductive window on the sidewall.
Two types of vertical coupling structures are utilized to achieve
the cross coupling between nonadjacent resonators at different
layers. With multilayer capability, there is more flexibility to
arrange the cavities of coupled resonator filters in 3-D space.
It is demonstrated by both the simulation and experiment that
the proposed filter structures occupy a compact circuit area and
have good selectivity. The filter with electric field cross coupling
occupies a half area of a planar four-pole waveguide filter, while
the filter with stacked vias cross coupling has 65% size reduction
in comparison with a planar waveguide filter.
Index Terms—Bandpass filter, cavity, coupling coefficient, low-
temperature co-fired ceramic (LTCC), stacked vias, quasi-elliptic
function.
I. INTRODUCTION
M
ODERN microwave communication systems require
high-performance bandpass filters with high selectivity,
low insertion loss, and compact size. Filters with a waveguide
structure can offer low loss and a high quality (
) factor, but
usually at the price of large size, heavy weight, and high cost.
The manufacturing of the waveguide also needs sufficient
accuracy in order to operate at the millimeter-wave frequency.
Recently, the concept of synthesized rectangular waveguide
structures [1] has attracted much interest. The waveguide is
dielectric filled and embedded into a substrate. The sidewall
of the rectangular waveguide can be realized by arrays of
metallic via or metallic grooves. This kind of waveguide not
only maintains a good
factor, but also suits the realization
of high-performance bandpass filters at the millimeter-wave
frequency regime.
Several direct-coupled cavity filters have been realized by
the synthesized waveguide structures on flip-chip modules [2],
printed circuit board [3], and thick-film technology [4]. These
filters usually occupy a large circuit area because of the planar
arrangement of the resonators. With the mature multilayer tech-
nology, synthesized waveguide filters are also fabricated on a
Manuscript received February 6, 2007; revised May 10, 2007. This work was
supported in part by the National Science Council, Taiwan, R.O.C., under Grant
NSC 93-2752-E-002-003-PAE and Grant NSC 94-2219-E-002-001 and by the
Industrial Technology Research Institute.
The authors are with the Department of Electrical Engineering and Grad-
uate Institute Communication Engineering, National Taiwan University, Taipei,
10617 Taiwan, R.O.C. (e-mail: rbwu@ew.ee.ntu.edu.tw).
Digital Object Identifier 10.1109/TMTT.2007.902080
low-temperature co-fired ceramic (LTCC) technology [5] and
micromachined process [6]. Multilayer filter technology pro-
vides significant benefits in terms of design flexibility and den-
sity. This makes vertical coupling between resonators possible
and cavities can be piled up in 3-D space, which will largely re-
duce the circuit area.
Frequency selectivity is also an essential feature of a high-
performance filter. Quasi-elliptic or elliptic filters will have
transmission zeros at finite frequencies and give more im-
proved stopband rejection than conventional direct-coupled
filters [7]–[9]. Such filter responses can be realized with cross
coupling between nonadjacent resonators [10]. The zeros are
then obtained by means of destructive interference between the
different signal path connecting the input and output ports [11].
Recently, a conventional parallel coupled microstrip filter with
a transmission line inserted inverter for realization of different
advanced filtering characteristics was presented in [12]. With
the additional cross-coupled transmission line, there is greater
flexibility in the arrangement of the cross coupling path to
achieve the desired frequency response.
In this paper, quasi-elliptic bandpass filters with a cross-cou-
pling architecture are developed in the multilayered LTCC tech-
nology, as shown in Figs. 1 and 2. An open-ended microstrip
line is used to excite the filters by a narrow slot etched on the
first/last cavity. The LTCC resonators can be stacked three-di-
mensionally to provide various coupling mechanisms required
in the design of quasi-elliptic bandpass filters, while achieving
compact sizes and good selectivity. The cross coupling between
nonadjacent resonators is achieved by a square aperture at the
center of the common wall in Fig. 1 or by additional stacked vias
and short-circuited coplanar waveguides (CPWs) in Fig. 2.
This paper is organized as follow. Section II describes the
key design parameters required to realize the quasi-elliptic
filter. Section III introduces several coupling structures and the
relation of coupling coefficients versus physical dimensions.
Section IV provides two design examples. The experiment data
are presented and compared with simulation results. Finally,
some brief conclusions are drawn in Section V.
II. F
ILTER DESIGN
A general coupling structure of a quasi-elliptic filter is de-
picted in Fig. 3 [10], where each node represents a resonator, and
the solid and dashed lines indicate the main and cross-coupling
paths, respectively. It is essential that the signs of the coupling
coefficients
and are opposite in order to
realize a pair of attenuation poles at the finite frequencies. This
means that the coupling routes of
and
0018-9480/$25.00 © 2007 IEEE
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1772 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 8, AUGUST 2007
Fig. 1. Structure of a four-pole quasi-elliptic waveguide filter in a multilayer
configuration. (a) 3–D overview. (b) Side view.
(a)
(b)
Fig. 2. Structure of a vertically stacked quasi-elliptic waveguide filter. (a) 3-D
overview. (b) Side view.
need to be out of phase. However, it does not matter which one
is positive or negative, as long as their signs are opposite.
Fig. 3. General coupling structure of a quasi-elliptic filter.
The design parameters of bandpass filters, i.e., the coupling
coefficients and the external
factor in Fig. 3, can be deter-
mined in terms of the circuit elements of a low-pass prototype
filter [13].
After determining the required coupling coefficients and ex-
ternal
factor, the relationship between coupling coefficients
and physical structures of coupled resonators should be estab-
lished in order to determine the physical dimensions of the filter
against the design parameters. The coupling coefficients of cou-
pled resonators can be specified by two split resonate frequen-
cies resulting from electromagnetic coupling [14], i.e.,
(1)
In (1),
and are defined to be the lower and higher res-
onance frequencies, respectively. The sign of the coupling co-
efficient is dependent on the physical structure of the coupled
resonators. For filter design, the meaning of positive or negative
coupling is rather relative. The positive and negative coupling
will have an opposite phase response, which can be found by
the
-parameter of the coupling structure.
The external
factor can be characterized by [13]
(2)
where
and represent the resonance frequency and the
3-dB bandwidth of the input or output resonator. By (1) and
(2), design curves of the coupling coefficients and external
factor versus physical dimensions of coupled resonators can be
established. The sizes of coupling structures are also obtained
according to the design parameters.
III. R
EALIZATION OF COUPLING
COEFFICIENTS
A. LTCC Cavity Resonator
The cavity resonator is formed by several stacked dielectric
substrate with metal surfaces at the outer layers and via arrays
as vertical sidewalls, which is shown in Fig. 4. The resonant
frequencies of the cavity with a perfectly conducting wall can
be obtained by [15]
(3)
where
is the relative dielectric constant, is the speed of light,
and are the width, height, and length of the cavity, respec-
tively, and
and are the indices of the resonant mode. By
(3), the initial dimensions of the synthesized waveguide cavity
can be determined, and the final values are optimized by the
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SHEN et al.: DESIGN OF VERTICALLY STACKED WAVEGUIDE FILTERS IN LTCC 1773
Fig. 4. Cavity resonator with metallic plates and via arrays.
Fig. 5. Field patterns of mode of a single rectangular cavity. (a) Electric
field. (b) Magnetic field. (c) Surface current on the metal plane.
eigenmode solution solver of a full-wave simulator, e.g., the
High Frequency Structure Simulator (HFSS). Besides, the
factor of a cavity resonator increases with the cavity height. To
get a higher
cavity, more substrate layers will be used to form
a synthesized cavity according to the fabrication limitation.
The cavity resonators in Figs. 1 and 2 operate in their funda-
mental mode
at a common center frequency. Fig. 5 shows
the field patterns. The electric field is mainly concentrated at
the center of the cavity and in the direction normal to the metal
plane. The magnetic field is tangential to the metallic walls and
rotates in the cavity. The magnetic field increases its strength
gradually when approaching the sidewalls. The surface current
will flow into the center of the metal plate in a radial shape.
Next, several structures of coupled resonators in LTCC tech-
nology are introduced. The cavities in the same layer are cou-
pled by an inductive window, while the cavities between dif-
ferent layers are coupled by a square aperture in the center of
Fig. 6. Coupling coefficients of the slot coupling structure.
the common wall, a narrow slot at the edge of the common wall,
or an additional thru via.
B. Magnetic Coupling by Broad-Wall Slots
To efficiently couple two adjacent cavities in different layers,
a narrow slot in the common intermediate wall is placed near
the sidewalls of the cavity and in the direction perpendicular to
the surface current. It will significantly interrupt the surface cur-
rent flow and introduce strong coupling, analogous to the design
principle of waveguide slot antenna [16]. Hence, the coupling
between adjacent cavities can be achieved by means of mag-
netic fields through the narrow slot in the common wall.
The coupling coefficient is affected by the length and position
of the narrow slot. To get strong direct coupling, the narrow
slot should be located as close to the sidewall of the cavity as
possible. The coupling strength is then controlled by the slot
length. Fig. 6 shows the relation between the slot length and
coupling coefficients of the stacked cavities. As mentioned in
Section II, the coupling coefficients are calculated by two split
resonate frequencies, which can be obtained by an eigenmode
solution solver of HFSS. Each cavity resonates at 31 GHz, with
the cavity size
mm and the
relative dielectric constant
. According to fabrication
limitation, the slot is located 0.2 mm from the cavity sidewall.
Due to the presence of the slot, the length of each cavity
should be adjusted to compensate for the shifted resonant fre-
quency [2].
will be the difference between the original cavity
length
and the modified cavity length for the frequency
compensation. The relation between the coupling coefficients
and the variation of the cavity length is also plotted in Fig. 6.
Both the adjustments in slot length and cavity length are nor-
malized to the cavity width
in Fig. 6.
C. Magnetic Coupling by Narrow-Wall Window
The pair of vias composing the inductive window are used
to control the coupling of the cavities at the same layer. It is a
common coupling structure in the planar waveguide filter [3].
The coupling strength is controlled by the separation of the via
pair. The wider the separation, the stronger the coupling can be.
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1774 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 8, AUGUST 2007
Fig. 7. Coupling coefficient of the inductive window coupling structure.
Fig. 8. Coupling coefficients of the square aperture coupling structure.
Fig. 7 shows the relation of coupling coefficients versus the via
pitch (VP) and the cavity length variation
.
D. Electric Coupling by Broad-Wall Aperture
If a square aperture is opened at the center of two stacked
cavities, where the electric field is a maximum, the coupling can
be achieved in terms of an electric field normal to the aperture.
Fig. 8 shows the aperture length (AL) and cavity length devi-
ation
, which are both normalized to cavity width , versus
the coupling coefficients.
E. Cross Coupling by Vias Connecting Nonadjacent Cavities
As shown in Fig. 2, the first and last cavities are coupled by an
additional through via, which provides the cross-coupling path
to achieving a quasi-elliptic frequency response [17]. The cross-
coupling structure is mainly formed by short-circuited CPW
feed lines with a main thru-hole via and two shorter buried vias
beside the main through via. The CPW feed lines are connected
to the first and last resonators and the main thru-hole via is con-
nected to the CPW feed lines.
In the LTCC process, the electric field of a grounded CPW is
mainly concentrated under the signal line because of the small
Fig. 9. Via coupling structure. (a) Overview. (b) Coupling coefficient.
substrate height and wide gap. The electric field distribution of
a grounded CPW will be similar to that of
of a cavity
resonator and, therefore, energy can be gathered from the cavity
easily with a CPW feed line. The energy will pass down along
the main thru via and couple to the other cavity connected to the
CPW feed line. Two shorter buried vias provide current return
paths when energy is delivering.
The coupling coefficient between the first and last cavities can
be extracted by a very weak excitation with the same method de-
scribed in [2]. Two split resonant frequencies can be seen clearly
from the
-parameter of the coupled resonators structure. The
strength of the cross coupling can be controlled by the length
of the CPW stretched into a cavity. When the short-circuited
end of the CPW is closer to the center of the cavity where the
electric field is strongest, more energy can be gathered from the
resonator. The relation between the coupling coefficient and the
CPW length is shown in Fig. 9. The coupling coefficient basi-
cally increases with the CPW length.
Fig. 10 plots the
-parameter of coupled resonators with a
slot coupling structure and a via coupling structure. By com-
paring the phase responses in Fig. 10(a) and (b), it is clear that
they are out-of-phase. That is to say, two extracted coupling co-
efficients have opposite signs [18]. Therefore, slot coupling and
via coupling structures can be used to realize a quasi-elliptic
filter.
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SHEN et al.: DESIGN OF VERTICALLY STACKED WAVEGUIDE FILTERS IN LTCC 1775
Fig. 10. Phase response of coupled resonators. (a) Slot coupling structure.
(b) Via coupling structure.
IV. DESIGN EXAMPLES AND EXPERIMENTAL VERIFICATION
Next, two kinds of cross coupling structures are utilized to
realize the four-pole quasi-elliptic filters. When designing the
filters, perfect conductor sidewalls are assumed for calculation
efficiency. Based on the above procedure, the initial dimensions
of the coupling slots, aperture, and inductive window are de-
cided. The cavities sizes with small variation are also known.
The entire filter structure is then optimized by HFSS to meet
the design specification. After the initial design of the filter is
accomplished, the metallic via arrays take the place of the per-
fect conductor sidewalls to complete the filter design.
A. Feeding Structure
The filter is excited by open-ended microstrip lines, as shown
in Fig. 1. The slot discontinuity at the return path of the mi-
crostrip line causes strong coupling for the same reason that
waveguide slot antenna radiates. To maximize the magnetic cou-
pling, a virtual short is placed at the center of each slot by using a
quarter-wavelength open stub beyond the slots center [19]. This
kind of feeding structure can not only avoid dc power loss, but
Fig. 11. External factor of the microstrip line feeding structure.
also contribute to fabrication simplicity. The external
factor
of the feeding structure is controlled by the external slot length
and position. Fig. 11 shows the relation of the external
factor
versus the slot length
and the cavity length variation .
B. Basic Stacked LTCC Filters Design
A canonical waveguide filter with coupling between nonadja-
cent cavities can be utilized to achieve an elliptic-function filter
response [20]. The cross couplings are achieved by a circle at
the center of the common wall or by a narrow slot at the edge
of the cavity. In the same concept, a quasi-elliptic filter realized
by LTCC technology is presented here.
The configuration of the basic stacked LTCC filter is shown
in Fig. 1. The coupling produced by means of electric and mag-
netic fields have opposite signs [21]; therefore, the filter archi-
tecture of Fig. 1 will conform to the general coupling structure
in Fig. 3, which results in a quasi-elliptic frequency response.
The four-pole quasi-elliptic waveguide filter is designed and
fabricated in LTCC. The specification of the filter is 10% frac-
tional bandwidth centered at 31 GHz with 20-dB passband re-
turn loss. The element values of the low-pass prototype filter
are found to be
and . By [13], the coupling coefficients
and I/O external
factor are
(4)
The relative dielectric constant of the substrate is 7.8 and its loss
tangent is 0.0078 at 30 GHz. The thickness of each metal layer
is 13
m and the dielectric layer thickness between two metal
layers is 50
m. The cavity height is 250 m, while the mi-
crostrip substrate height is 150
m. The via diameter is 100 m.
To allow on-wafer measurement by coplanar probes, the input
and output probe pad should be on the same layer. Therefore, a
vertical transition composed of thru-hole vias is utilized to con-
nect the bottom microstrip line to the top layer. Eight grounded
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1776 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 8, AUGUST 2007
Fig. 12. (a) Hole structure of a four-pole quasi-elliptic waveguide filter with a
vertical transition. (b) Fabricated filter.
Fig. 13. Geometric parameters of the quasi-elliptic filter.
vias are located around the thru-hole via to mimic a coaxial
transmission line effect.
Fig. 12(a) shows the whole filter configuration and Fig. 12(b)
is a photograph of the fabricated filter. Geometric parameters
of the filter are illustrated in Fig. 13 and summarized in Table I.
The overall size of the four-pole quasi-elliptic LTCC waveguide
filter without the vertical transition is 4.46
2.72 0.8 mm ,
i.e., approximately
, where is the
guided wavelength on the substrate at the center frequency. A
full-wave simulator HFSS is used to calculate the
factor of a
cavity. The
factor is found to be approximately 103, which is
similar to the measured data of approximately 99.
The frequency response of the filter is shown in Fig. 14, where
the solid and dashed lines denote the measured and simulated re-
sults, respectively. The dashed–dotted lines represent the ideal
TABLE I
G
EOMETRIC
PARAMETERS OF
THE QUASI-ELLIPTIC
FILTER
Fig. 14. Simulation and measurement results of the four-pole quasi-elliptic
LTCC bandpass filter.
circuit response. The simulation result is not fully identical with
the theoretical response. It can be contributed to the replacement
of vertical sidewalls of the cavities by via arrays in LTCC and the
vertical transitions for on-wafer measurement. When the perfect
sidewalls are substituted by the via arrays, the major difference
is the in-band return loss. This may be contributed to the varia-
tions in the coupling coefficients and external
factors, which
make the frequency response deviated from that by the theo-
retical one. When the vertical transition is taken into consider-
ation, The major discrepancy is the deterioration at the higher
frequency side of the passband.
The measured center frequency of the filter is 30.9 GHz and
the 3-dB bandwidth is 3.85 GHz. The passband insertion loss is
approximately 2.55 dB and the passband return loss is greater
than 12 dB. Two attenuation poles near the cutoff frequencies of
the passband can be clearly identified. The two attenuation poles
are located at 28.2 and 34 GHz. The measured results are in good
agreement with the full-wave simulation results by HFSS.
C. Fully Stacked LTCC Filters Design
The filter in Fig. 2 introduces a novel structure composed
of vertically stacked cavities to realize a quasi-elliptic function
filter. The configuration is re-plotted in Fig. 15, composed of
four vertically stacked synthesized rectangular cavities. Adja-
cent cavities are coupled to each other by a narrow slot near the
edge of the common wall. The cross-coupling path is realized
by short-circuited CPW feed lines connected to the first and last
resonators with a main thru-hole via connection.
A vertically stacked four-pole quasi-elliptic waveguide filter
is designed and fabricated by the same LTCC process in the pre-
vious design example. The dimensions of other coupling slots
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SHEN et al.: DESIGN OF VERTICALLY STACKED WAVEGUIDE FILTERS IN LTCC 1777
Fig. 15. Layer sketch of a fully stacked quasi-elliptic waveguide filter without
via arrays.
Fig. 16. (a) Entire structure of a fully stacked four-pole quasi-elliptic wave-
guide filter with a vertical transition. (b) Fabricated filter.
and cavities can be determined under the same design guide.
Here, the specification of the filter is 10% fractional bandwidth
centered at 30.2 GHz with 20-dB passband return loss. Same el-
ement values of the low-pass prototype filter in the previous ex-
ample are used. The coupling coefficients and external
factor
are equal to (4) because the same fractional bandwidth is chosen.
The whole filter configuration and the photograph of the fab-
ricated filter are shown in Fig. 16. The cavity height is 150
m.
The microstrip line substrate height is 100
m. To simplify
the measurement, the input microstrip line at the top layer will
feed the filter from the opposite direction. A vertical transi-
tion connecting microstrip lines at the top and bottom layers is
used for on-wafer measurement, as mentioned in the previous
example. Geometric parameters of the filter are illustrated in
Fig. 17. Geometric parameters of the vertically stacked quasi-elliptic filter.
(a) Top view of the first and fourth cavity. (b) Top view of the second and third
cavity.
TABLE II
G
EOMETRIC
PARAMETERS OF THE VERTICALLY
STACKED QUASI-ELLIPTIC FILTER
Fig. 17 and summarized in Table II. The size of the vertically
stacked four-pole quasi-elliptic LTCC waveguide filter without
the vertical transition is 3.67
2.4 0.8 mm , i.e., approxi-
mately
.
The frequency response of the filter is shown in Fig. 18, where
the solid and dashed lines denote measured and simulated re-
sults, respectively. The dashed–dotted lines represent the ideal
circuit response. The measured center frequency of the filter is
29.5 GHz and the 3-dB bandwidth is 3.93 GHz. The passband
insertion loss is approximately 2.8 dB and the passband return
loss is greater than 12 dB. The two attenuation poles are located
at 26.85 and 33.05 GHz. The measured center frequency has
down shifted approximately 2% as compared to the simulation
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1778 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 8, AUGUST 2007
Fig. 18. Simulation and measurement results of the vertically stacked four-pole
quasi-elliptic LTCC bandpass filter.
result. This may be contributed to the smaller LTCC shrinkage
due to more metal plates used in this filter configuration. There-
fore, the cavities are bigger than expected and will result in the
down-shifted center frequency.
V. C
ONCLUSION
New structures have been proposed to realize the various cou-
pling mechanisms required for quasi-elliptic bandpass filters de-
sign using stacked LTCC cavities. The idea has been validated
by presenting two four-pole quasi-elliptic function bandpass fil-
ters in LTCC. Several coupling mechanisms between adjacent
and nonadjacent resonators have been described in details. By
this multilayer technology, the vertical coupling between cavi-
ties at different layers can be achieved and the filters will have
compact size as compared to the conventional planar filters.
The filter with the electric field cross-coupling structure ap-
proximately occupies the size of two cavities, while the footprint
of the filter with fully stacked cavities and cross-coupling via
structure can achieve nearly 65% size reduction as compared to
the conventional planar four-pole waveguide filters. The cross
coupling between nonadjacent resonators is introduced to ex-
hibit a single pair of transmission zeros near the passband at
finite frequencies and, thus, much better selectivity. As a result,
the proposed structures of the filters occupy a compact circuit
area and have a good stopband response.
A
CKNOWLEDGMENT
The authors would like to thank Dr. H.-H. Lin, C.-L. Wang,
and C.-C. Chuang, all with the Computer and Communica-
tion Laboratory, Institute of Technology Industrial Research,
Hsinchu, Taiwan, R.O.C., for their help in the fabrication and
measurement of the LTCC filters.
R
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Tze-Min Shen was born in Chiayi, Taiwan, R.O.C.,
on August 5, 1981. He received the B.S. degree in
electrical engineering and M.S. degree in communi-
cation engineering from National Taiwan University,
Taipei, Taiwan, R.O.C., in 2004 and 2006, respec-
tively, and is currently working toward the Ph.D.
degree in communication engineering at National
Taiwan University.
His research interests is the design of microwave
filters.
Authorized licensed use limited to: National Taiwan University. Downloaded on February 25, 2009 at 01:50 from IEEE Xplore. Restrictions apply.
SHEN et al.: DESIGN OF VERTICALLY STACKED WAVEGUIDE FILTERS IN LTCC 1779
Chi-Feng Chen was born in PingTung, Taiwan,
R.O.C., on September 3, 1979. He received the
B.S. degree in physics from Chung Yuan Christian
University, Taoyuan, Taiwan, R.O.C., in 2001, the
M.S. degree in electrophysics from National Chiao
Tung University, Hsinchu, Taiwan, R.O.C., in 2003,
and the Ph.D. degree in communication engineering
from National Taiwan University, Taipei, Taiwan,
R.O.C., in 2006.
His research interests include the design of
microwave filters and associated RF modules for
microwave and millimeter-wave applications.
Ting-Yi Huang was born in Hualien, Taiwan,
R.O.C., on November 12, 1977. He received the B.S.
degree in electrical engineering and M.S. degree in
communication engineering from National Taiwan
University, Taipei, Taiwan, R.O.C., in 2000 and
2002, respectively, and is currently working toward
the Ph.D. degree in communication engineering at
National Taiwan University.
His research interests include computational elec-
tromagnetics, the design of microwave filters, transi-
tions, and associated RF modules for microwave and
millimeter-wave applications.
Ruey-Beei Wu (M’91–SM’97) was born in Tainan,
Taiwan, R.O.C., on October 27, 1957. He received
the B.S.E.E. and Ph.D. degrees from National Taiwan
University, Taipei, Taiwan, R.O.C., in 1979 and 1985,
respectively.
In 1982, he joined the faculty of the Department of
Electrical Engineering, National Taiwan University,
where he is currently a Professor. He is also with the
Graduate Institute of Communications Engineering,
which was established in 1997. From March 1986 to
February 1987, he was a Visiting Scholar with IBM,
East Fishkill, NY. From August 1994 to July 1995, he was with the Electrical
Engineering Department, University of California at Los Angeles. He was ap-
pointed the Director of the National Center for High-Performance Computing
from May 1998 to April 2000 and the Directorate General of Planning and Eval-
uation Division from November 2002 to July 2004, both under the National Sci-
ence Council. Since August 2005, he has been Chairperson of the Department
of Electrical Engineering, National Taiwan University. He has authored or coau-
thored over 150 papers in international journals or conferences. He served as an
Associate Editor of the
Journal of Chinese Institute of Electrical Engineering in
1996. His research interests include computational electromagnetics, transmis-
sion line and waveguide discontinuities, microwave and millimeter-wave planar
circuits, and interconnection modeling for computer packaging.
Dr. Wu is a member Phi Tau Phi and the Chinese Institute of Electrical
Engineers. He has been an associate editor for the IEEE T
RANSACTIONS ON
MICROWAVE
THEORY AND
TECHNIQUES
since 2005. He is an elected Executive
Committee member of the IEEE Microwave Theory and Techniques Society
(IEEE MTT-S) Taipei Chapter. He is an elected Executive Committee member
of the Institute of United Radio Science (URSI) Taipei Section. He was the
recipient of the Distinguished Research Award presented by the National
Science Council (1990, 1993, 1995, and 1997) and the Outstanding Electrical
Engineering Professor Award presented by the Chinese Institute of Electrical
Engineers (1999).
Authorized licensed use limited to: National Taiwan University. Downloaded on February 25, 2009 at 01:50 from IEEE Xplore. Restrictions apply.
