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12 PRZEGL D ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 88 NR 1a/2012
Andrei BLINOV1, Dmitri VINNIKOV1, Volodymyr IVAKHNO2, Volodymyr ZAMARUEV2
Tallinn University of Technology (1), Kharkiv Polytechnical Institute (2)
Hybrid IGBT-IGCT Switch
Abstract. This paper presents an analysis of a hybrid high-voltage switch based on the parallel connection of IGBT and IGCT. The proposed
configuration allows combining the advantages of both semiconductors, resulting in substantially reduced power losses. Such energy efficient
switches could be used in high-power systems where decreased cooling system requirements are a major concern. The operation principle of the
switch is described and simulated and power dissipation is estimated at different operation conditions.
Streszczenie. Artyku prezentuje analiz! hybrydowego "cznika wysokonapi!ciowego bazuj"cego na równoleg ym po "czeniu tranzystora IGBT
i tranzystora IGCT. Proponowana konfiguracja pozwala na uzyskanie zalet dwóch pó przewodników, w rezultacie czego otrzymano znaczne
zmniejszenie strat mocy. Takie energooszcz!dne "czniki, mog" by# u$ywane w systemach du$ej mocy, gdzie zmniejszenie urz"dze% systemu
ch odzenia jest g ównym problemem. Opisano i zasymulowano dzia anie "czników, oraz oszacowano rozpraszanie energii dla ró$nych warunków
pracy. ( !cznik hybrydowy typu IGBT-IGCT).
Keywords: Insulated gate bipolar transistors, insulated gate-commutated thyristors, industrial power systems.
S owa kluczowe: Tranzystor bipolarny z izolowan bramk , tyrystor z izolowan bramk , Przemys!owe systemy energetyczne.
Introduction
High power densities together with a high functionality
are the key aspects of modern power electronics. Further
requirements are decreased volume and weight of the
power systems as well as low cost. In order to fulfil these
demands high switching frequencies of the semiconductors
are necessary. Insulated gate bipolar transistors (IGBTs)
are the major representatives in present day’s medium- and
high voltage electronics. In terms of blocking voltages (up to
6.5 kV) these devices have reached a level which can
satisfy the majority of needs. The major advantages of
IGBTs are easy driving and snubberless operation [1]. On
the other hand, the switching behaviour of low voltage class
IGBTs (<1 kV) is generally slower in comparison to
MOSFETs, and high voltage class IGBTs (>3.3 kV)
generally have higher conduction losses than GTO and
IGCTs. In order to improve the performance of IGBTs,
different approaches and methods were introduced and
developed. For instance, at lower voltages, increased
performance was achieved by a parallel IGBT-MOSFET-
combination as shown in [2]. The hybrid integration of a
unipolar and a bipolar power semiconductor in parallel
allowed combining of their advantages whilst avoiding their
disadvantages [3]. However, these positive results were
observed only for certain applications and operation
parameters.
Similarly, for high power applications the performance of
high-power switches could be increased by a parallel
connection of IGBT and IGCT switches [4-6]. This paper will
focus on 4.5 kV class switches, since both IGBT and IGCT
type semiconductors in press-pack type housings are
commercially available, allowing easy connection of these
devices in series by special cooling systems. The rated
permanent DC voltage for both semiconductor devices is
generally 2.8 kV. Using two- or three-level topologies, if
necessary, this is sufficient to cope with the requirements of
many traction and industrial applications with voltage
ratings of 2.0-5.6 kV without the need of series connection
of several semiconductors. Comparing two 4.5 kV class
press-pack semiconductors: T0900EA45A- Westcode
(Table 1 [7]) and 5SHY35L4512- ABB (Table 2 [8]), it could
be observed that the on-state voltage UT of IGCT is lower
than the corresponding parameter UCE(sat) of IGBT. The
turn-on behaviour is similar for both devices, while turn-off
behaviour of IGCT is distinctly slower, which results in
greatly increased losses during turn-off.
The idea is based on the integration of positive
properties of gate-commutated thyristors in terms of low
turn-on and on-state power losses as well as high surge
current capability and IGBTs with their relatively low losses
during turn-off. This may allow creating high-voltage and
high-current energy-efficient switches with increased
switching frequency, which could be advantageous in high-
power (>500 KVA) industrial and railway traction
systems [9].
Operation principle
The structure of the proposed hybrid switch (HS)
configuration is presented in Fig. 1. The HS consists of a
parallel connected asymmetrical press-pack IGCT and
press-pack IGBT with an integrated freewheeling diode
(FWD).
In the following analysis the HS is assumed to be
operated in voltage-source inverter (VSI) circuits. The test
circuit shown in Fig. 2 represents the main events that could
occur in VSI topologies and includes the clamp circuit, HS,
D1 (representing FWD of the opposite HS) and inductive
load. The inductances LCL and LD represent the stray
inductance of the clamp and the stray inductance between
the IGCT and IGBT housings, respectively. The clamp
circuit typically used in IGCT applications limits the surge
reverse-recovery current of the turning-off FWD and
generally consists of a di/dt limiting inductor Li, a clamp
capacitor CCL, a clamping diode DCL and a resistor RS.
Table 1. Characteristic values of 900A 4.5kV IGBT (T0900EA45A)
Parameter Symbol Value
Collector-emitter voltage UC
E
4500 V
Permanent DC voltage UDC 2800 V
Collector-emitter saturation voltage
(IC=900 A) UCE(sat) 4.7 V
Turn-on delay time td
(
on
)
1.6 µs
Rise time tr 2.3 µs
Critical rate of rise of diode current dIt/dtcr 2000 A/µs
Turn-off delay time td
(
off
)
1.2 µs
Fall time tf 1.2 µs
Turn-off energy (IC=900 A) Eoff 2.6 J
Table 2. Characteristic values of 4.0kA 4.5kV IGCT(5SHY35L4512)
Parameter Symbol Value
Peak off-state voltage UDRM 4500 V
Permanent DC voltage UDC 2800 V
On-state voltage (I
T
=900 A) U
T
1.15 V
Turn-on delay time td
(
on
)
3.5µs
Rise time tr 1 µs
Critical rate of rise of current dIt/dtcr 1000 A/µs
Turn-off delay time td
(
off
)
11 µs
Turn-off energy (I
T
=900 A) Eoff 6-8 J
PRZEGL D ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 88 NR 1a/2012 13
Fig.1. Proposed hybrid switch configuration
Fig.2. Configuration of the simulation circuit
The generalised HS operation principle is shown in
Fig. 3 and the following time intervals during the operation
period can be distinguished:
t0 " the beginning of each switching period of PWM. The
thyristor SA of the HS is turned on by the control signal,
applying full load current. During this time the transistor of
the HS is turned off.
t0-t1 " freewheeling diode reverse-recovery process,
duration and behaviour are dependent on the diode type
and di/dt.
t1-t2 " thyristor is conducting with low losses. The
voltage across the HS determined by the voltage drop
across the thyristor UT.
t2 " the turn-off control impulse is applied to the thyristor
and simultaneously the turn-on impulse is applied to the
transistor SB of the HS.
t2-t3 " as the turn-on behaviour of the IGBT is faster
than the turn-off transient of the IGCT, the thyristor turn-off
process occurs when the transistor is already in the on-
state. The load current is distributed between both
semiconductors.
t3 " the SA returns to the blocking state, the full load
current is applied to the transistor SB. Hence, the turn-off
transient of the thyristor occurs when the voltage is limited
to the voltage drop UCE(sat) across the conducting transistor
SB of the HS. Moreover, during the current transfer to the
transistor the voltage across its terminals is limited to the
voltage drop across the SA during the on-state. The
required duration of the transistor on-state should not be
shorter than the turn-off transient of the thyristor.
t4 " the turn-off of the HS occurs by applying negative
gate voltage to the transistor after the thyristor returns to the
blocking state. The turn off transient of the HV IGBTs is
generally 2…7 µs. After the transistor is switched off, the
voltage across HS and all its components become equal to
the supply voltage.
Simulation model
To simulate the HS operation the commutation circuit
shown in Fig. 2 was modelled in PSpice software. The
same diode model was used in the topology for simplicity
and the following simulation parameters were assumed: the
input voltage is 2800 V, the maximum load current is 750 A.
The values of the circuit’s passive components are
determined according to [10].
Fig.3. Generalised operation principle and switching waveforms of
the proposed HS
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I (A)
U (V)
Time (us)
56,7*3 86,*+3 86,*43
I (SA) I (SB)
U (HS)
Fig.4. Simulated turn-off behaviour of HS at I=750 A, UDC=2800 V
The simulations confirm the estimated behaviour of the
proposed switch configuration. At turn-on the HS operates
like an IGCT with the di/dt clamp, ensuring that D1 is
operating within its SOA. The on-state voltage of the HS is
equal to the voltage drop across the thyristor during its
conducting period. During turn-off of the HS the transistor is
turned on for a short period; the turning-off thyristor current
is then transferred to the transistor, which is closed right
after the thyristor current becomes zero. The turn-off
dynamics of the HS are greatly increased, while the
excellent on-state characteristics of IGCT remain (Fig. 4)
and all the elements are operated within the SOA.
Generalised loss evaluation
In the simulations of losses, the minimum IGBT
switching losses with a very small gate resistances of
RGon=4 # and RGoff=2.5 # are assumed. In real industrial
converters the IGBT gate units are adjusted to generate the
14 PRZEGL D ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 88 NR 1a/2012
desired di/dt and dU/dt to avoid large voltage and current
spikes during transients. However, the use of the gate
resistor to control the di/dt results in substantially higher
switching losses in IGBT [11]. If a di/dt limiting turn-on
snubber is used with both IGBT and IGCT devices, the turn-
on losses would be similar [12]. On the other hand, the turn-
off losses of the device may increase slightly [13].
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!
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&
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'
'% "( %) (# )$ $
Switching frequen cy (Hz)
Switch Current (A)
89+: 89;: 7*
Fig.5. Switch switching frequency vs. current for different
semiconductor configurations corresponding to 3 kW total power
dissipation at UDC=2800 V, D=0.5
!
&
'
"
%
(
&% '% "% %% (% )% #% $% ! %
Average losses (W)
Switching freque ncy (Hz)
89+: 89;: 7*
Fig.6. Switch power dissipation vs. switching frequency for different
semiconductor configurations at I=750 A, UDC=2800 V, D=0.5
The turn-off losses of the IGCT were excluded in the
simulations; however, according to the test results
presented in the previous papers [14], the turn-off losses
may not be completely removed due to several factors.
Firstly, for a large area device, such as the IGCT, a
significant output capacitance must be charged in order to
establish the depletion region to support voltage. Another
factor is the free carriers which had not recombined being
swept from the junction. Nevertheless, an 89% reduction in
turn-off losses was reported in [15]. In real conditions, the
power losses of industrial applications could be distinctly
higher than the simulated values.
After the turn-on of the IGBT, the current distribution
between conducting IGBT and IGCT is mainly influenced by
different characteristics of the semiconductors, temperature
differences and asymmetrically distributed stray
inductances in the circuit [16][17]. Assuming both
semiconductors in conducting state, the current sharing
inside the HS neglecting cell resistances and inductances
can be calculated by
(1) )(
)(
)(
IU
IU
I
I
k
T
satCE
SB
SA
I
where ISA and ISB are IGCT and IGBT currents respectively
!%
&%
'%
"%
%%
(%
<! <& <' <" <% <( <) <# <$
Average losses (W)
Duty Cycle
89+: 89;: 7*
Fig.7. Switch power dissipation vs. duty cycle for different
semiconductor configurations at I=750 A, UDC=2800 V, fsw=750 Hz
$ )% ( "% '%
%
!
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'
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Switch current
Average losses (W)
Switching freque ncy (Hz)
89;: 89+:
Fig.8. Breakdown of power losses of HS corresponding to 3 kW
total power dissipation at UDC=2800 V, D=0.5
Using Eq. (1) the IGBT and IGCT currents could be
obtained by
(2)
I
SB k
II
!
"
1
1
(3)
I
I
SA k
k
II
!
"
1
According to simulations shown of Figs. 5-8 (IGBT-
T0900EA45A; IGCT - 5SHY35L4512), the IGCT is showing
better dynamics for currents above 650 A, whereas the
IGBT is performing better at lower currents (Fig. 5). The
proposed switch configuration is estimated to provide
2.3…2.8 times increased switching frequency in
comparison to single hard switched IGBT or IGCT exhibiting
the same power dissipation of 3 kW. Assuming the same
switching frequency in the range of 250…1050 Hz and
switch current of 750 A the IGBT performs better than IGCT
at frequencies above 450 Hz, whereas the HS provides
substantial (1.9…2 times) decrease in power losses in
comparison to single semiconductors (Fig. 6). Fig. 7 shows
average losses of all considered switch solutions operating
in the studied circuit with the wide range of duty cycles. The
IGBT performs better than IGCT up to D=0.85. Again, the
HS shows substantially (1.8…2.2 times) reduced power
dissipation in comparison to single semiconductors. As
Fig. 8 reveals, the losses of transistor in the HS are
essentially (2.4…4.4 times) higher than the losses of
thyristor under considered operation parameters. The loss
distribution becomes more equal with an increased switch
current.
PRZEGL D ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 88 NR 1a/2012 15
Fig.9. Comparison of semiconductor prices of studied switch
configurations
Unlike in the case of the typical parallel connection of
identical semiconductors, in the proposed HS both switches
are conducting full input current during the operation, thus
the current rating of both semiconductors must be sufficient.
On the other hand, the overall power dissipation is
decreased in comparison with single switches allowing one
to increase the switching frequency or reduce cooling
system requirements. Moreover, if one of the
semiconductors fails, the other one can still continue to
operate independently unless sufficient cooling is applied.
The economical feasibility of the HS implementation
greatly depends on the application and its operation
conditions. The comparison of semiconductor prices of
discussed switch configurations is shown in Fig. 9. It should
be mentioned, that semiconductor price is only a part of the
overall power electronic system. The prices of the passive
components greatly vary for different applications and are
not considered in this paper.
Conclusions
Using commercially available 4.5 kV class IGBTs and
IGCTs in press-pack housings it is possible to create
energy efficient switches with essentially decreased power
losses. Despite having decreased maximum current
capabilities in comparison with parallel connected identical
transistors or thyristors and a higher price than single
semiconductor switches, the proposed switch configuration
could be beneficial in applications where higher switching
frequencies are required or decreased cooling system
requirements are essential.
Acknowledgement
This research work has been supported by Estonian
Ministry of Education and Research (Project
SF0140016s11), Estonian Science Foundation
(Grant ETF7572) and Estonian Archimedes Foundation
(project „Doctoral school of energy and geotechnology II“).
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Authors: M.Sc. Andrei Blinov, E-mail: andrei.blinov@ieee.org and
Dr.Sc.techn. Dmitri Vinnikov, E-mail: dmitri.vinnikov@ieee.org –
Tallinn University of Technology, Department of Electrical Drives
and Power Electronics, Ehitajate tee 5, Tallinn, 19086, Estonia.
Prof. Volodymyr Ivakhno, E-mail: v-ivakhno@ukr.net and Prof.
Volodymyr Zamaruev E-mail: vvz@kpi.kharkov.ua – Kharkiv
Polytechnical Institute, Department of Physical and Biomedical
Electronics, Frunze 21, Kharkiv, 61002, Ukraine.
Przegląd Elektrotechniczny
2012 | R. 88, nr 1a | 12-15
