Content uploaded by Ashraf Uz Zaman
Author content
All content in this area was uploaded by Ashraf Uz Zaman on Sep 05, 2014
Content may be subject to copyright.
2992 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 6, JUNE 2014
Wide-Band Slot Antenna Arrays With Single-Layer
Corporate-Feed Network in Ridge Gap
Waveguide Technology
Ashraf Uz Zaman and Per-Simon Kildal, Fellow, IEEE
Abstract—Single-layer, wideband, and low-loss corporate-feed
networks for slot antenna arrays are described. The antenna is
built using ridge gap waveguide technology, formed between two
parallel metal plates without the requirements of electrical con-
tact between these plates. The corporate-feed network is realized
by a texture of pins and a guiding ridge in the bottom plate, and
the radiating slots are placed in the smooth top plate. The paper
describes two test antennas: a 4 1 linear slot array and a 2 2
planar slot array. Both have been fabricated and tested at Ku-
band. The linear array shows more than 20% bandwidth and the
22 array shows a bandwidth of 21% for 10-dB return loss. There
are good agreements between measured and simulated patterns for
both antennas. Measured gain for the planar array is found to be
at least 12.2 dBi over 12–15 GHz band.
Index Terms—Corporate-feed, perfect electrical conductor
(PEC), perfect magnetic conductor (PMC), ridge gap waveguide,
waveguide slot array antenna, wideband.
I. INTRODUCTION
PLANAR-ARRAY antennas are suitable for a lot of ap-
plications requiring high to moderate antenna gain. Mi-
crostrip antenna arrays and waveguide slot arrays are the two
main planar antenna technologies, which have been used ex-
tensively over a wide range of frequencies. Microstrip arrays
are compact, easy to manufacture, cost-effective and easy to
integrate with active electronics. However, the microstrip feed
networks suffer from high ohmic and dielectric losses at high
frequency [1], [2]. Spurious radiations and leakage in the form
of surface waves are always major concerns in microstrip an-
tennas and are difficult to handle [3]. All these lead to sub-
stantial reduction in gain and antenna efficiency. Substrate in-
tegrated waveguide (SIW) or post-wall based planar array an-
tennas have been proposed to realize low cost solutions [4], [5].
This technology enables integration of active circuits together
withtheantennas.ThelossesinSIWarebetterthanmicrostrip
Manuscript received September 03, 2013; revised February 05, 2014; ac-
cepted February 21, 2014. Date of publication March 05, 2014; date of current
version May 29, 2014. This work was supported in part by the Swedish Re-
search Council VR, and by The Swedish Governmental Agency for Innovation
Systems (VINNOVA) within the VINN Excellence Center Chase.
The authors are with the Department of Signals and Systems, Chalmers
University of Technology, SE-412 96 Göteborg, Sweden (e-mail:
zaman@chalmers.se; per-simon.kildal@chalmers.se).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2014.2309970
and coplanar structures. Still, losses may be of concern, espe-
cially for high-gain (above 30 dBi) antennas, due to the pres-
ence of dielectric material [6], [7]. On the other hand, wave-
guide slot arrays have low losses. They suffer neither from di-
electric nor radiation loss. So, waveguide slot arrays are con-
sidered for many applications requiring high gain and high effi-
ciency. However, wideband waveguide slot arrays require cor-
porate-feed networks that become very complex and bulky. At
high frequencies, such networks require accurate and high pre-
cision (and thereby expensive) manufacturing so as to achieve
good electrical contacts between the slotted metal plate and the
bottom feed structure [8]. The present paper will describe a com-
pact ridge gap waveguide corporate-feed network without the
need of metal contact between these two plates, thereby having
simple mechanical assembly and potentially lower manufac-
turing cost.
Apart from the manufacturing costs and assembly compli-
cations, some other limitations of waveguide slot arrays have
been reported in literature. Series-fed single layer waveguide
slot arrays are simple but have narrow bandwidth due to the
long line effect [9] (i.e., different delays to each element). In a
single-layer structure, it is normally not possible to feed each ra-
diating element in parallel (full corporate-feed) because of the
space limitations associated with keeping the element spacing
smaller than one wavelength to avoid grating lobe [10],
[11]. Therefore, there is needed a complex multilayer feed net-
work. Such a double-layer corporate-feed network in rectan-
gular waveguide technology is described in [12]. It was real-
ized with diffusion bonding of laminated thin metal plates and
the antenna worked over 11% relative bandwidth. The present
paper describes a much simpler and single-layer corporate-feed
network.
Based on the above discussion, existing corporate-feed net-
work technologies have limitations with respect to bandwidth
and mechanical simplicity, giving an opportunity for the pro-
posed gap waveguide technology. The gap waveguide tech-
nology introduced in [13] uses the basic cutoff of a perfect
electrical conductor–perfect magnetic conductor (PEC-PMC)
parallel-plate waveguide configuration to control desired elec-
tromagnetic propagation between the two parallel plates. In
the realized gap waveguide, the PEC-PMC cutoff becomes
a stopband, and this stopband can be achieved by a periodic
structure, such as metal pins or mushrooms in the textured sur-
faces [14]. However, the textured surface must also incorporate
guiding structures in the form of ridges, grooves or strips. As
0018-926X © 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
ZAMAN AND KILDAL: WIDE-BAND SLOT ANTENNA ARRAYS WITH SINGLE-LAYER CORPORATE-FEED NETWORK 2993
a result of the stopband, the electromagnetic waves can prop-
agate along these ridges, grooves or strips without leaking
away in other directions. These guiding structures thereby
define three different gap waveguides or transmission lines,
referred to as ridge, groove and microstrip gap waveguides
[15]. Thus, gap waveguide technology offers mechanically
flexible guiding structures where good electrical contact be-
tween the building blocks is not an issue at all. Consequently,
high-precision metal machining required for assembling and
manufacturing waveguide slot arrays can be avoided and cost
of production can be minimized. The first experimental valida-
tions of the ridge gap waveguide and the suspended microstrip
gap waveguide were published in [16]–[18], respectively. In
the present paper, we will use ridge gap waveguide to de-
sign a linear slot array and a planar slot array. The ridge gap
waveguide structure used in the linear array design is shown
in Fig. 1(a). The planar array antenna, on the other hand, uses
periodic pin structure with a bit different dimension. The stop-
band and the dispersion diagram for both the pin structures
are discussed in Section II.
Until now, there have been no complete papers on gap
waveguide antennas, except for some conference papers de-
scribing horn antenna [19], initial works in single hard-wall
waveguide slot array in [20], and single slot design in ridge
gap waveguide [21]. We should also mention the multi-layer
phased array antenna and dual mode horn array based on related
technology presented in [22], [23]. Recently, series fed groove
gap waveguide slot array having inclined slots in narrow wall
has also been reported [24], but this slot array was narrow band
(5% BW). In the present paper, we document wideband 4 1
element linear slot array and a 2 2 element planar slot array
designed at Ku band based on ridge gap waveguide technology.
Both of these antennas are designed for a fixed broadside beam
and a 20% relative bandwidth. In a wideband antenna, the
reflection coefficient of each component constituting the
antenna must be very good to avoid interference maxima of the
total of the complete antenna over all the frequencies within
the band of interest. The wideband single slot element and a
wideband single T-junction based on ridge gapwaveguide were
first presented in [21]. A similar slot element and T-junction
have been used in the linear array in the present paper. This
is explained in Section IV. However, the present 2 2planar
slot antenna has more compact feed-network geometry, and the
shape of the slot and the feed T-section is different. The 2 2
planar antenna design is described in Section V. The linear
array is excited with a conventional coaxial SMA connector
with an extended center conductor, and the 2 2planararray
is excited with a transition from ridge gap to rectangular wave-
guide. Measured results for both these antennas are presented
in Section VI and are carefully compared with the simulated
results.
The design of the two slot arrays in the paper has been done
entirely by numerical simulations using a cut-and-try approach
based on intelligent guessing from previous experience. The
ridge gap waveguide modes look similar to the modes along in-
verted microstrip lines, so the topologies of different ridge gap
waveguide circuits will look similar to those of microstrip cir-
cuits [25], which is of help in the design process. The topology
Fig. 1. (a) Detailed dimensions of the periodic metal pin and ridge gap wave-
guide geometry used in this paper. (b) Dispersion diagram of the unit cell of
the periodic pins used in the linear array design, mm, mm,
mm, and mm. (c) Dispersion diagram of the unit cell of the
periodicpinsusedintheplanararraydesign, mm, mm,
mm, and mm. (d) Simulated for different rows of
pins around the guiding ridge.
of ridge gap waveguides is too complex for analytic treatment,
although some analytic works have been done [26], [27].
2994 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 6, JUNE 2014
II. DESIGNING THE STOPBAND
As mentioned in [16], the main performance of the gap wave-
guide is determined by its ability to create parallel-plate stop-
band for wave propagation in undesired directions. This ability
is determined by the quasi-periodic pin structure and the height
of the air gap. The size of the stopband is directly linked to the
operational bandwidth of the power dividers and feeding net-
work, and hence to the whole gap waveguide antenna. The stop-
band study consists of parametric sweeps of all parameters asso-
ciated with the periodic structure to be used (a, p, d, and in
this paper), and it is based on Eigen mode analysis of the single
unit cell by enforcing periodic boundary conditions on the sides
of the structure. The details of such types of studies can be found
in [14].
For both the slot arrays in the present paper we use a tex-
tured surface made of square pins, designed to have the stopband
in Ku-band, covering 12–15 GHz. The dispersion diagrams for
the parallel-plate geometry with metal pins that are used in the
linear array and the 2 2planararrayareshowninFig.1(b)
and (c), respectively. The pin dimensions for the two cases are
also shown in the figures.
As shown in the above two figures, a large stop-band is cre-
ated by the pin surface after 10 GHz where all the parallel-plate
modes are in cutoff. Once the dimensions of the periodic pin
structure are obtained, the ridge can be incorporated within the
periodic pin structure. The ridge height is kept same as the pin
height everywhere in the gap waveguide structures. It is to be
noted that- while designing the bends of the T-junctions, a few
pins have to be removed or relocated locally. As long as the
modification is done locally over one or two pins, this does not
have severe consequence on the designed stop-band and on the
performance of the gap waveguide structures, provided the re-
locations are done locally over one or two pins.
Apart from the stopband design, the number of pin rows
around the guiding ridge is also very important for gap wave-
guide antennas as it will dictate the element spacing of the
antenna and grating lobe issue. So, we have done a simple
two-port analysis of the basic ridge gap waveguide structure
with varying number of pin rows. We have simulated a 150 mm
(at 15 GHz) ridge gap waveguide with three, two and
single rows of pins, respectively. The is almost identical for
the three and two rows of pins. For single row of pins, the is
degraded by 0.1 dB over the whole Ku-band. This is shown in
Fig. 1(d). Thus, two pin rows are preferable for such structures
considering losses and isolation an issue. For designing the an-
tennas without grating-lobes, we have relaxed this requirement
and used one row of pin at some locations to have the adjacent
elements in less than spacing. We have done a tradeoff and
expect some leakage to neighboring elements but this leaked
energy will be as low as 20 dB after one row of pins [13] and
is acceptable for many antenna designs.
III. DESIGN RULES FOR THE SINGLE ELEMENT,POWER
DIVIDER,AND TRANSITIONS
We have designed both antennas based on full wave simula-
tions. First, we have determined the geometry of the half-wave-
length resonant slot element fed by a simple ridge gap wave-
guide. Then, we have tried to improve the bandwidth by adding
Fig. 2. (a) Drawing of the slot element with T-section; mm,
mm, mm, mm, mm, mm,
and mm. The air gap between pins and the top plate is 1 mm, even though
it is shown bigger in the drawing. (b) Simulated for the slot element with
and without the T-shape in the feeding ridge section. (c) Variation in parameter
of the T-shape section, is kept 0.5. (d) Variation in parameter
of the slot element, is kept 8.15 mm.
a T-section to the feeding ridge structure under the slot element
showninFig.2(a).Thelargestbandwidthisobtainedwhenthe
first resonance from the slot itself and the second resonance
from the feeding T-section appeared close to each other. These
ZAMAN AND KILDAL: WIDE-BAND SLOT ANTENNA ARRAYS WITH SINGLE-LAYER CORPORATE-FEED NETWORK 2995
two resonance deeps are clearly seen in the simulated plots
in Fig. 2(b).
Parametric sweeps have been done for several design param-
eters such as and and (shown in Fig. 2) to
achieve reasonable bandwidth of the single slot element. During
the parameter sweeps, we have found that the parameters
and have larger impact on the impedance bandwidth. On the
other hand the parameters such as and have relatively
smaller effect on the reflection coefficient of the single slot ele-
ment. The influence in simulated due to variations in
and are shown in Fig. 2(c) and (d).
The power dividers are based on well-known two-way
power division technique (T-junctions) used in microstrip
technology. The power divider for the linear array case is
based on impedance matching of the input line using a single
quarter-wavelength section which gives 30% bandwidth. The
power divider used in the 2 2 planar array case is based on a
gradual tapering from low-impedance to high impedance. No
transformers were used and thereby it could be made more
compact in size, and the bandwidth obtained was still 22%
which is enough for our demonstrator.
The stepped transition [Fig. 5(a)] from ridge gap waveguide
to groove gap waveguide used in the linear array case is based
on using different sections of Chebyshev transformers for the
impedance matching. This technique is well described in [28].
The 2 2 planar array has a 90 transition from ridge gap wave-
guide to rectangular waveguide (Fig. 9). The starting point of
this transition design is the well-known microstrip to waveguide
transition in [29]. As the mode in ridge gap waveguide is also
a quasi-TEM mode, ridge gap waveguide to rectangular wave-
guide transition can be designed in a similar way as the mi-
crostrip transition. Later, we have optimized the ridge gap wave-
guide transition to work well within our frequency of interest.
IV. LINEAR ARRAY DESIGN
Initial simulated results for the Ku-band linear slot array were
presented in [30]. The slot element used in that work was exited
with a ridge having the width mm. That ridge had
a prolonged T-shaped section that was added to achieve good
impedance bandwidth. The results presented in [30] were done
with slot elements having sharp rectangular corners. However,
slots with sharp corners are difficult to manufacture and need
very small diameter tool to mill the slot. To address this issue,
the slot element was redesigned with rounded corners. The new
slot element with rounded corners is shown in previous section.
The simulated reflection coefficient for this new slot element
is shown in Fig. 3(a). Apparently, there is almost no effect of
the rounded corners compared to sharp corners if the dimension
of the slot is retuned. The dimensions of the T-section are kept
unchanged. The dimension of the slot element is chosen in such
a way that the slot length to width ratio “ ” is kept smaller
than 0.5. This ratio is important for the suppression of the cross-
polarization. The thickness “ ” of the top metal plate is chosen
to be 2 mm.
The four-way power divider needed to build the feeding net-
work is based on the T-junction presented in [21]. In the initial
design, the two output lines of the T-junction were separated by
Fig. 3. (a) Simulated for the rounded edge slot element. (b) Four-way
power divider with the T-junctions; mm, mm. The upper
plate is shown lifted up and partly removed for clarity. (c) Simulated results for
the four-way power divider.
two rows of pins. For the array antenna, we have a limitation
in terms of spacing between the adjacent elements. With two
rows of pins, it is not possible to have the separation between
2996 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 6, JUNE 2014
Fig. 4. Complete linear array geometry. The upper plate is shown lifted and
partly removed for clarity.
the adjacent elements smaller than . Therefore, we designed
a more compact power divider with a four-way power division.
The compact four-way power divider is shown in Fig. 3(b) and
the simulated performance for this power divider is shown in
Fig. 3(c). The simulated S-parameter results demonstrate good
performance. Over the frequency band 11–15 GHz, equal power
distribution is achieved and the input reflection coefficient
is lower than 20 dB.
The linear array has been designed for a fixed broadside
beam. Therefore, the spacing between the elements was chosen
to be to ensure no grating lobes. All the slot elements
are excited with equal amplitude and phase, and the array was
designed to operate from 12 to 14 GHz. The complete array
antenna is shown in Fig. 4. Fig. 4 also shows a transition from
ridge gap waveguide to groove gap waveguide.
The propagating mode in a groove gap waveguide is very
similar to the mode of rectangular waveguide [25]. There-
fore, in the groove gap waveguide part the linear array antenna
can be excited in an easy way by a coaxial SMA connector
with an extended center conductor and a usual back short. The
details of the ridge gap to groove gap transition are shown in
Fig. 5(a). Simulated results for this transition and the complete
antenna are shown in Fig. 5(b). The designed stepped transition
is very wideband and the reflection coefficient of the tran-
sition is not the main contributing factor in of the complete
antenna. Instead, the of the slot element and the four-way
power divider dominates and adds up in phase after 15 GHz.
This causes the overall of the antenna to rise above 10
dB around 15 GHz. This 4 1 element linear array will have a
directive symmetric radiation pattern in H-plane. The E-plane
pattern will be wide, and this will cause significant back radi-
ation because the slots are close to the edge. The level of the
back radiation has been reduced by adding 2 corrugations with
quarter wavelength depth at the lowest frequency of operation.
The simulated radiation patterns for this linear array at 13 GHz
are shown in Fig. 6.
V. P LANAR 22ARRAY DESIGN
In a two-dimensional array, it is more difficult to design the
corporate-feed network because the element spacing must be
Fig. 5. (a) Stepped transition from ridge gap waveguide to groove gap wave-
guide, mm, mm, mm, mm,
mm; mm, mm (top metal plate not shown). (b)
Simulated for the stepped transition from ridge gap waveguide to groove
gap waveguide and the complete linear array antenna.
Fig. 6. Simulated radiation patterns for the linear array at 13 GHz.
smaller than one wavelength in two orthogonal directions. For
this reason, the feed network must be much more compact. Also,
the slot elements must be redesigned. The slot element with the
new dimensions is shown in Fig. 7(a) and the simulated reflec-
tion coefficient for this new slot element is shown in Fig. 7(b).
The ‘ ’ of the new slot element has been changed to 0.55
instead of 0.49 of the previous case. Also, the length of the
T-section “ ” has been changed from 8.15 mm to 6.85 mm
ZAMAN AND KILDAL: WIDE-BAND SLOT ANTENNA ARRAYS WITH SINGLE-LAYER CORPORATE-FEED NETWORK 2997
Fig. 7. (a) Schematic of the modified slot element; mm,
mm, mm, ,mmand mm. The upper plate is
shown lifted. (b) Simulated reflection coefficient for the modified slot element.
in this new design. The width of the exiting ridge has also been
changed from mm to mm.
The compact four-way power divider is designed without
using a transformer section. Instead, the widths of the
two divided ridge sections are gradually tapered to make the
impedance match. This is shown in Fig. 8(a) and the simulation
results for this four-way power divider are shown in Fig. 8(b).
This new power divider is not as wideband as the one in
Section III. Still, the power divider works well with below
20 dB and equal power division from 12 GHz to 15 GHz.
After designing the new slot element and compact power
divider, a simple 2 2 element planar array was designed to
operate over the band 12–15 GHz. The 2 2elementarrayis
excited in phase and with equal amplitude by a corporate feed
network. The element spacing is chosen to be 17.5 mm, which
is about at 15 GHz.
This antenna is excited with a standard Ku-band rectangular
waveguide from the bottom plane using a transition from
Fig. 8. (a) Schematic of the modified T-junctions and four-way power divider,
mm ,mmand mm. (b) Simulated S-parameters
for the compact four-way power divider.
Fig. 9. Rectangular waveguide to ridge gap waveguide transition,
mm; mm; mm; mm;
mm.
rectangular waveguide to ridge gap waveguide. This transition
is shown in Fig. 9 and the complete antenna is shown in
Fig. 10(a). The simulated reflection coefficient for the antenna
with the transition and the single transition performance is
shown in Fig. 10(b). The simulated radiation pattern for this
22elementplanararrayat15GHzisshowninFig.11.
2998 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 6, JUNE 2014
Fig. 10. (a) Complete 2 2 element array geometry; element spacing is
17.5 mm. (b) Simulated for the bottom transition from ridge gap wave-
guide to rectangular waveguide and the complete planar antenna.
Fig. 11. Simulated E-plane and H-plane radiation patterns for the planar array
at 15 GHz.
VI. MEASUREMENT RESULTS
A. Reflection Coefficients
Figs. 12 and 13 shows the manufactured linear array antenna
and the planar antenna, respectively. Both these antennas are
made in aluminum by metal milling technique with tolerance
Fig. 12. Manufactured prototype of 4 1 element linear array antenna.
Fig. 13. Manufactured prototype of 2 2 element planar array antenna.
Fig. 14. Measured and simulated for the 4 1 element linear array
antenna.
Fig. 15. Measured and simulated for the 2 2 element planar array an-
tenna.
in the order of 25–30 m. The input reflection coefficients,
are shown in Figs. 14 and 15 for the 4 1 element linear array
and the 2 2 element planar antenna, respectively. It is apparent
ZAMAN AND KILDAL: WIDE-BAND SLOT ANTENNA ARRAYS WITH SINGLE-LAYER CORPORATE-FEED NETWORK 2999
Fig. 16. (a) Measured H plane radiation patterns for the linear array. (b) Mea-
sured E plane radiation patterns for the linear array.
from Figs. 14 and 15 that good input matching is achieved over
a wide range of frequencies (more than 20% relative bandwidth)
for both these manufactured antenna prototypes. Also, the mea-
sured parameters are in good agreement with the simulated
values. For the linear array, the difference in the level of the
S-parameters and the little frequency shift can be attributed to
variation of the length of the center conductor of the SMA probe,
which was cut manually at the lab. In case of the 2 2planar
array, there are additional peaks in the measured param-
eter. In the simulation, the 2 2 planar array was excited with
a waveguide port at the rectangular waveguide opening. In re-
ality, when the antenna is measured with a waveguide flange,
there is an additional transition from rectangular waveguide to
SMA probe. This extra transition is assumed to cause the addi-
tional peaks in the measured parameterofthe2 2planar
antenna.
B. Radiation Patterns
The radiation characteristics of both these antennas were
measured in our own anechoic chamber. The measured ra-
Fig. 17. (a) Measured H plane radiation patterns for the planar array. (b) Mea-
sured E plane radiation patterns for the planar array. (c) Measured cross-polar-
ization level for the planar array.
diation patterns are shown in Fig. 16(a) and (b) at several
frequency points in two principal planes of the linear array. In
Fig. 17(a) and (b), the measured patterns in the two principal
planes of the 2 2 array are also shown.
3000 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 6, JUNE 2014
Fig. 18. Simulated directivity and measured gain for the planar array.
For the 2 2 array, the cross-polarization in the E-plane
has been measured at two frequency points and presented in
Fig. 17(c). As presented in Figs. 16 and 17, the measured and
simulated radiation patterns are in quite good agreements for
both these two antennas. For the linear array antenna, the first
sidelobe level in H-plane is below 11 dB over the whole
frequency band of interest. For the 2 2 array, the sidelobe
is below 12.5 dB in H-plane for all the measured frequency
points. In E-plane, the sidelobe levels remain below 10 dB at
the lower frequencies, but they rise up to a level of 7.65 dB
at the upper end of the frequency band. These high sidelobes
are also present with a level of 8.35 dB in simulated patterns.
They are caused by the omnidirectional pattern of each indi-
vidual slot in E-plane and the large element spacing (close to
at higher frequency). Also, the array has an extent limited to
only two elements in each direction, so we do not have a clear
effect of the array factor. For a larger array, this first sidelobe
will appear closer to the main lobe and will become lower.
For the present array it is combined with the part of the wide
grating lobe that appears in real space along the ground plane
at 90 from broadside. The cross-polarization level in E-plane
is measured to be 24 dB, but this level is only related to the
accuracy and the misalignments in the measurement chamber.
C. Measured Gain
The realized gain of the 2 2 element planar array is also mea-
sured. The simulated directivity and measured gain versus fre-
quency are shown in Fig. 18. The dotted lines show the max-
imum available directivity of an aperture having dimension of
(35 35) mm (i.e., twice the element spacing in both planes)
and when the aperture efficiency is 75%. This is a correct refer-
ence aperture of a 2 2 array [31]. The maximum directivity is
calculated by the known formula , valid for big apertures
or big arrays. For the present case of 2 2 element small array,
it is difficult to achieve high aperture efficiency. The measured
gain is found varying from 12.2 to 13.8 dB within the frequency
range 12–15 GHz. The difference between simulated directivity
and realized gain is mainly due to the mismatch loss, the losses
in feeding network. The ohmic losses in feeding network are
found to be less than 0.2 dB over the entire band of interest.
VII. CONCLUSION
Novel single-layer corporate-feed networks for slot array
antennas based on the planar gap waveguide technology has
been proposed in this work. A 4 1 element linear slot array
and 2 2 element array have been designed, manufactured and
measured. The mechanical assemblies of both antennas are very
robust and flexible. The mechanical construction has no require-
ments at all for good electrical contact between the radiating slot
layer and the feed network layer. The radiating slot element, the
feeding network and the transitions used in both the antennas
have been designed to suppress the reflection coefficient and the
results show more than 20% relative bandwidth of the whole
arrays. Achieved impedance bandwidth is much larger than for
other arrays of similar size described in [12], [32], and [33].
The measured and simulated reflection coefficients agree quite
well for both antennas. The measured radiation patterns are also
in reasonable agreement with the computed patterns. The mea-
suredgainforthe2 2 element planar array is found to be at least
12.2 dBi. Thus, the proposed antenna exhibits promising features
such as planar geometry, low ohmic losses and wide bandwidth.
The presented 2 2 slot array element can be used as a sub-
array to build up much larger antenna. The element spacing of
17.5 mm corresponds to , which is small enough to avoid
grating lobes. The 2 2 array occupies only a unit cell area of
35 35 mm including the distribution network. This small ex-
tent has been possible by compressing the distribution network.
Therefore, we have used only one pin row between some parts
of the distribution network, instead of the recommended two
rows. Still, the distribution network worked. In order to use the
22 array as a sub-array in a large array it must be optimized
numerically as a unit cell in an infinite array configuration using
periodic boundary conditions, so as to include the effects of mu-
tual coupling between sub-arrays. The feeding network for the
full array can be designed by using the same 3 dB power dividers
that we have used here, in a full corporate distribution network
connecting all sub-arrays. The present work shows that it should
be possible to use ridge gap waveguides to realize high gain, low
loss slot arrays consisting of two metal plates, one with a texture
of ridges and pins, and the other with an array of slots, and with
an air gap between the two plates. However, some parts of the
ridge on the textured plate are quite thin which may complicate
the manufacturing at higher frequencies.
ACKNOWLEDGMENT
The authors would like to thank Dr. Ali Khalegi of K. N.
Toosi University of Technology, Iran for his help regarding the
manufacturing of the antennas.
REFERENCES
[1] A. Borji, D. Busuioc, and S. Safavi-Naeini, “Efficient, low-cost inte-
grated waveguide-fed planar antenna array for Ku-band applications,”
IEEE Antennas Wireless Propag. Lett., vol. 8, pp. 336–339, 2009.
[2] D. M. Pozar, “Considerations for millimeter wave printed antennas,”
IEEE Trans. Antennas and Propagation, vol. 31, no. 5, pp. 740–747,
Sept. 1983.
[3] E. Levine, G. Malamud, S. Shtrikman, and D. Treves, “A study of mi-
crostrip array antennas with the feed network,” IEEE Trans. Antennas
Propag., vol. 37, no. 4, pp. 426–464, Apr. 1989.
[4] X.P.Chen,K.Wu,L.Han,andF.H.,“Low-costhighgainplanar
antenna array for 60-GHz band applications,” IEEE Trans. Antennas
Propag., vol. 58, no. 6, pp. 2126–2129, Jun. 2010.
ZAMAN AND KILDAL: WIDE-BAND SLOT ANTENNA ARRAYS WITH SINGLE-LAYER CORPORATE-FEED NETWORK 3001
[5] M.Bozzi,A.Georgiadis,andK.Wu, “Review of substrate-integrated
waveguide circuits and antennas,” IET Microw., Antennas Propag.,
vol. 5, no. 8, pp. 909–920, Jun. 2011.
[6] M.Bozzi,M.Pasian,L.Perregrini,andK.Wu,“Onthelossesinsub-
strate integrated waveguides and cavities,” Int. J. Microw. Wireless
Tech., vol. 5, no. 1, pp. 395–401, 2009.
[7] J. Hirokawa and M. Ando, “Efficiency of 76-GHz post-wall wave-
guide-fed parallel-plate slot arrays,” IEEE Trans. Antenna Propag.,
vol. 48, no. 11, pp. 1742–1745, Nov. 2000.
[8] Y.Kimura,T.Hirano,J.Hirokawa,andM.Ando,“Alternating-phase
fed single-layer slotted waveguide arrays with chokes dispensing with
narrow wall contacts,” in IEE Proc.-Microw. Antennas Propag,Oct.
2001, vol. 148, no. 5.
[9] S. Park, Y. Tsunemitsu, J. Hirokawa, and M. Ando, “Center feed single
layer slotted waveguide array,” IEEE Trans. Antennas Propag., vol. 54,
no. 5, pp. 1474–1480, May 2006.
[10] S. Fujii, Y. Tsunemitsu, G. Yoshida, N. Goto, M. Zhang, J. Hirokawa,
and M. Ando, “A wideband single-layer slotted waveguide array with
an embedded partially corporate feed,” in Proc. Int. Symp. Antennas
Propag., TP-C27-5,Oct.2008.
[11] T. Tsugawa, Y. Sugio, and Y. Yamada, “Circularly polarized dielec-
tric- loaded planar antenna excited by the parallel feeding waveguide
network,” IEEE Trans. Broadcasting, vol. 43, no. 2, pp. 205–212, Jun.
1997.
[12] Y. Miura, J. Hirokawa, M. Ando, Y. Shibuya, and G. Yoshida,
“Double-layer full-corporate-feed hollow-waveguide slot array an-
tenna in the 60 GHz-band,” IEEE Trans. Antennas Propag., vol. 59,
no. 8, pp. 2844–2851, Aug. 2011.
[13] P.-S.Kildal,E.Alfonso,A.Valero-Nogueira,andE.Rajo-Igles
ias,
“Local metamaterial-based waveguides in gaps between parallel metal
plates,” IEEE Antennas Wireless Propag. Lett., vol. 8, pp. 84–87, 2009.
[14] E. Rajo-Iglesias and P.-S. Kildal, “Numerical studies of bandwidth of
parallel plate cut-off realized by bed of nails, corrugations and mush-
room-type EBG for use in gap waveguides,” IET Microw., Antennas
Propag., vol. 5, no. 3, pp. 282–289, Mar. 2011.
[15] P.-S. Kildal, “Three metamaterial-based gap waveguides between par-
allel metal plates for mm/submm waves,” in Proc. 3rd Eur. Conf. An-
tennas Propag.,Berlin,Mar.2009.
[16] P.-S.Kildal,A.U.Zaman,E.Rajo-Iglesias,E.Alfonso,andA.Valero-
Nogueira, “Design and experimental verification of ridge gap waveg-
uides in bed of nails for parallel plate mode suppression,” IETMicrow.,
Antennas Propag., vol. 5, no. 3, pp. 262–270, Mar. 2011.
[17] A. U. Zaman, P.-S. Kildal, M. Ferndahl, and A. Kishk, “Validation of
ridge gap waveguide performance using in-house TRL calibration kit,”
in Proc. 4th Eur. Conf. AntennasPropag., Barcelona, Spain, Apr. 2010.
[18] A. Valero-Nogueira, J. Domenech, M. Baquero, J. I. Herranz, E. Al-
fonso, and A. Vila, “Gap waveguides using a suspended stripona
bed of nails,” IEEE Antennas Wireless Propag. Letters, vol. 10, pp.
1006–1009, 2011.
[19] A. Kishk and P.-S. Kildal, “Quasi-TEM H-plane horns with wideband
open hard side-walls,” in Proc. 4th Eur. Conf. Antennas Propag.,
Barcelona, Spain, Apr. 12–16, 2010.
[20] A. Valero-Nogueira, E. Alfonso, J. I. Herranz, and M. Baquero, “Planar
slot array antenna fed by oversized Quasi TEM waveguide,” Microw.
Opt. Technol. Lett., vol. 49, no. 8, pp. 1875–1877, 2007.
[21] A. U. Zaman and P.-S. Kildal, “Slot antenna in ridge gap waveguide
technology,” in Proc. 6th Eur. Conf. Antennas Propag., Prague, Czech
Republic, Mar. 2012.
[22] H. Kirino and K. Ogawa, “A 76 GHz multi-layered phased array an-
tenna using a non-metal contact metama terial waveguide,” IEEE Trans.
Antennas Propag., vol. 60, no. 2, pp. 840–853, Feb. 2012.
[23] E. Pucci, E. Rajo-Iglesias, J.-L. Vasquuez-Roy, and P.-S. Kildal,
“Planar dual-mode horn array with corporate-feed network in inverted
microstrip gap waveguide,” IEEE Trans. Antennas Propag., Jul. 2013,
to be published.
[24] S. M. Giner, A. Valero-Nogueira, J. I. H. Gerruzo, and M. Baquero,
“Excitation of untitled narrow wall slot in groove gap waveguide by
using a parasitic dipole,” in Proc. 7th Eur. Conf. Antennas Propag.,
Gothenburg, Sweden, Apr. 2013, pp. 3082–3085.
[25] H. Raza, J. Yang, P.-S. Kildal, and E. Alfonso, “Resemblance between
gap waveguides and hollow waveguides,” IET Microw., Antennas,
Propag., vol. 7, no. 15, pp. 1221–1227, Dec. 2013.
[26] M. Bosiljevac, Z. Sipus, and P.-S. Kildal, “Construction of Green’s
functions of parallel plates with periodic text ure with application to gap
waveguides—A plane wave spectral domain approach,” IET Microw.
Antennas Propag., vol. 4, no. 11, pp. 1799–1810, Nov. 2010.
[27] A. Polemi, S. Maci, and P.-S. Kildal, “Dispersion characteristics of
metamaterial-based parallel-plate ridge gap waveguide realized by bed
of nails,” IEEE Trans. Antennas Propag., vol. 59, no. 3, pp. 904–913,
Mar. 2011.
[28] J. Uher, J. Bornemann, and U. Rosenburg, Waveguide Components
for Antenna Feed Systems: Theory and Cad. Norwood, MA, USA:
Artech House, 1993, ISBN 0-89006-582-9.
[29] Y. C. Leong and S. Weinreb, “Full band waveguide-to-microstrip probe
transitions,” in IEEE Microw. Theory Tech. Symp. Dig., Jun. 1999, pp.
1435–1438.
[30] A.U.ZamanandP.-S.Kildal,“Ku-band linear slot-array in ridge gap-
waveguide technology,” in Proc. 7th Eur. Conf. Antennas Propag.,
Gothenburg, Sweden, Apr. 2013.
[31] P.-S. Kildal, “Fundamental directivity and efficiency limitations of
single and multi-port antennas,” in Proc. 2nd Eur. Conf. Antennas
Propag., Edinburgh, U.K., Nov. 2007.
[32] Y. Zhang, Z. N. Chen, X. Qing, and W. Hong, “Wideband millimeter-
wave substrate integrated waveguide slotted narrow-wall fed cavity an-
tennas,” IEEE Trans. Antennas Propag., vol. 59, no. 5, pp. 1488–1496,
May 2011.
[33] M. H. Waida and A. E. Fathy, “Substrate integrated waveguide
Ku-band cavity backed 2 2 microstrip patch array antenna,” IEEE
Antennas Wireless Propag. Lett., vol. 8, pp. 1054–1056, 2009.
Ashraf Uz Zaman was born in Chittagong,
Bangladesh. He received the B.Sc. degree in elec-
trical and electronics engineering from Chittagong
University of Engineering and Technology, Chit-
tagong, Bangladesh. In 2007 and the M.Sc. degree
from the Chalmers University of Technology,
Gothenburg, Sweden. He is currently working
toward the Ph.D. degree with the Communica-
tion, Information Theory, and Antenna Division,
Chalmers University of Technology.
His main research interest includes millimeter and
sub-millimeter waveguide technology, frequency-selective surfaces, microwave
passive components, packaging techniques, integration of MMIC with the an-
tennas, etc.
Per-Simon Kildal (M’76–SM’81–F’95) received
two doctoral degrees from the Norwegian Institute
of Technology, Trondheim, Norway.
He has been a Professor in antennas at Chalmers
University of Technology, Gothenburg, Sweden,
since 1989. He is heading the Antenna group.
His main tasks are to lead and supervise research
and education within antenna systems. Until now,
18 graduate students have received a Ph.D. from
him. He has done the electrical design of the 40
m120 m cylindrical reflector antenna and line
feed of the EISCAT scientific organization, and the dual-reflector Gregorian
feed of the 300-m radio telescope in Arecibo. He is the inventor behind
technologies such as dipole with beam forming ring, the hat antenna, and the
eleven feed. He was the first to introduce the reverberation chamber as an
accurate measurement instrument tool for over-the-air (OTA) characterization
of small antennas and wireless terminals for use in multipath environments
with fading. He is also the originator of the concept of soft and hard surfaces
in 1988, today being regarded as the first metamaterials concept. This concept
is the basis of his latest and most fundamental invention, the gap waveguide
technology. His research is innovative and industrially oriented, and has
resulted in several patents and related spinoff companies, the most known
being Bluetest AB. He organizes and lectures in courses within the European
School of Antennas (ESoA). His textbook Foundations of Antennas—A Unified
Approach (Studentlitteratur, 2000) was well received, and is now in the process
of being revised. He has authored more than 120 articles in scientific journals,
concerning antenna theory, analysis, design, and measurements.
Prof. Kildal was awarded best paper awards by the IEEE (1985 R.W.P. King
Award and 1991Schelkunoff Prize Paper Award) for two of his articles. In 2011,
he received the prestigious Distinguished Achievements Award from the IEEE
Antennas and Propagation Society.
