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Energy-Efficient High-Voltage Switch
Based on Parallel Connection of IGBT and IGCT
Andrei Blinov1, Dmitri Vinnikov1, Volodymyr Ivakhno2
1Tallinn University of Technology (Estonia), 2Kharkiv Polytechnical Institute (Ukraine)
andrei.blinov@ieee.org
Abstract- This paper presents an analysis of a hybrid high-
voltage switch based on the parallel connection of IGBT and
IGCT. The proposed configuration allows combining the
advantages of both semiconductors, resulting in substantially
reduced power losses. Such energy efficient switches could be
used in high-power systems where decreased cooling system
requirements are a major concern. The operation principle of
the switch is described and simulated and power dissipation is
estimated at different operation conditions.
I. INTRODUCTION
High power densities together with a high functionality are
the key aspects of modern power electronics. Further
requirements are decreased volume and weight of the power
systems as well as low cost. In order to fulfil these demands
high switching frequencies of the semiconductors are
necessary. Insulated gate bipolar transistors (IGBTs) are the
major representatives in present day’s medium- and high
voltage electronics. In terms of blocking voltages (up to
6.5 kV) these devices have reached a level which can satisfy
the majority of needs. The major advantages of IGBTs are
easy driving and snubberless operation [1]. On the other
hand, the switching behaviour of low voltage class IGBTs
(<1 kV) is generally slower in comparison to MOSFETs, and
high voltage class IGBTs (>3.3 kV) generally have higher
conduction losses than GTO and IGCTs. In order to improve
the performance of IGBTs, different approaches and methods
were introduced and developed. For instance, at lower
voltages, increased performance was achieved by a parallel
IGBT-MOSFET-combination as shown in [2]. The hybrid
integration of a unipolar and a bipolar power semiconductor
in parallel allowed combining of their advantages whilst
avoiding their disadvantages [3]. However, these positive
results were observed only for certain applications and
operation parameters.
Similarly, for high power applications the performance of
high-power switches could be increased by a parallel
connection of IGBT and IGCT switches [4-6]. This paper will
focus on 4.5 kV class switches, since both IGBT and IGCT
type semiconductors in press-pack type housings are
commercially available, allowing easy connection of these
devices in series by special cooling systems. The rated
permanent DC voltage for both semiconductor devices is
generally 2.8 kV. Using two- or three-level topologies, if
necessary, this is sufficient to cope with the requirements of
many traction and industrial applications with voltage ratings
of 2.0-5.6 kV without the need of series connection of several
semiconductors. Comparing parameters of two 4.5 kV class
press-pack semiconductors: T0900EA45A-Westcode
(Table 1 [7]) and 5SHY35L4512-ABB (Table 2 [8]), it could
be observed that the on-state voltage UT of IGCT is lower
than the corresponding parameter UCE(sat) of IGBT. The turn-
on behaviour is similar for both devices, while turn-off
behaviour of IGCT is distinctly slower, which results in
greatly increased losses during turn-off (Fig. 1).
The idea is based on the integration of positive properties
of gate-commutated thyristors in terms of low turn-on and on-
state power losses as well as high surge current capability and
IGBTs with their relatively low losses during turn-off. This
may allow creating high-voltage and high-current energy-
efficient switches with increased switching frequency, which
could be advantageous in high-power (>500 KVA) industrial
and railway traction systems.
TABLE I
CHARACTERISTIC VALUES OF 900 A 4500 V IGBT (T0900EA45A)
Para meter Symbol Value
Collector-emitter voltage UCE 4500 V
Permanent DC voltage U
DC
2800 V
Collector-emitter saturation voltage (I
C
=900 A) UCE
(
sat
)
4.7 V
Turn-on delay time td(on) 1.6 µs
Rise time t
r
2.3 µs
Critical rate of rise of diode current dIt/dtc
r
2000 A/µs
Turn-off delay time td(o
ff
) 1.2 µs
Fall time t
f
1.2 µs
Turn-off energy (I
C
=900 A) Eo
ff
2.6 J
TABLE II
CHARACTERISTIC VALUES OF 4000 A 4500 V IGCT (5SHY35L4512)
Para meter Symbol Value
Peak off-state voltage U
D
RM 4500 V
Permanent DC voltage U
DC
2800 V
On-state voltage (I
T
=900 A) U
T
1.15 V
Turn-on delay time td(on) 3.5µs
Rise time t
r
1 µs
Critical rate of rise of current dIt/dtc
r
1000 A/µs
Turn-off delay time td(o
ff
) 11 µs
Turn-off energy (I
T
=900 A) Eo
ff
6-8 J
II. OPERATION PRINCIPLE
The structure of the proposed hybrid switch (HS)
configuration is presented in Fig. 2. The HS consists of a
parallel connected asymmetrical press-pack IGCT and press-
pack IGBT with an integrated freewheeling diode (FWD).
In the following analysis the HS is assumed to be operated
in voltage - source inverter (VSI) circuits. The test circuit
shown in Fig. 3 represents the main events that could occur in
978-1-4244-8807-0/11/$26.00 ©2011 IEEE
360
0
0.5
1
1.5
2
2.5
3
3.5
4
350 580 810 1040 1270 1500
Switch Current (A)
0
1
2
3
4
5
6
7
IGBT Eoff
IGBT Uce(sat)
0
2
4
6
8
10
12
350 580 810 1040 1270 1500
Switch Current (A)
0
0.2
0.4
0.6
0.8
1
1.2
1.4
IGCT Eoff
IGCT UT
Estimated
Fig. 1. Side-by side comparison ofT0900EA45A IGBT and 5SHY35L4512
IGCT on-state voltages and turn-off energies vs. current
Fig. 2. Proposed hybrid switch configuration
Fig. 3. Configuration of the commutation circuit
VSI topologies and includes the clamp circuit, hybrid switch,
D1 (representing FWD of the opposite HS) and inductive
load. The inductances LCL and LD represent the stray
inductance of the clamp and the stray inductance between the
IGCT and IGBT housings, respectively. The values of these
inductances should be minimized in order to meet the
specified SOA of the devices. The clamp circuit typically
used in IGCT applications limits the surge reverse-recovery
current of the turning-off freewheeling diodes and generally
consists of a dI/dt limiting inductor Li, a clamp capacitor CCL,
a clamping diode DCL and a resistor RS. In the case of a
failure, the clamp inductance limits the short circuit current as
well.
t0t1t2t3t4
t
t
t
t
t
t
Control
SA
I
Control
SB
U
SA
I
SB
I
I
HS
U
IN
U
ton
SB
on
t
T
UCE(sat)
U
Fig. 4. Generalised operation principle and switching waveforms of the
proposed HS
The generalised HS operation principle is shown in Fig. 4
and the following time intervals during the operation period
can be distinguished:
t0 − the beginning of each switching period of PWM. The
thyristor SA of the HS is turned on by the control
signal, applying full load current. During this time the
transistor of the HS is turned off.
t0-t1 − freewheeling diode reverse-recovery process, duration
and behaviour are dependent on the diode type and
dI/dt.
t1-t2 − thyristor is conducting with low losses. The voltage
across the HS determined by the voltage drop across
the thyristor UT.
t2 − the turn-off control impulse is applied to the thyristor
and simultaneously the turn-on impulse is applied to
the transistor SB of the HS.
t2-t3 − as the turn-on behaviour of the IGBT is faster than the
turn-off transient of the IGCT, the thyristor turn-off
process occurs when the transistor is already in the on-
state. The load current is distributed between both
semiconductors.
t3 − the SA returns to the blocking state, the full load
current is applied to the transistor SB. Hence, the turn-
off transient of the thyristor occurs when the voltage is
limited to the voltage drop UCE(sat) across the
conducting transistor SB of the HS. Moreover, during
the current transfer to the transistor the voltage across
its terminals is limited to the voltage drop across the
SA during the on-state. The required duration of the
361
transistor on-state should not be shorter than the turn-
off transient of the thyristor.
t4 − the turn-off of the HS occurs by applying negative gate
voltage to the transistor after the thyristor returns to
the blocking state. The turn off transient of the HV
IGBTs is generally 2…7 µs. After the transistor is
switched off, the voltage across HS and all its
components become equal to the supply voltage.
III. SIMULATION MODEL
A. Simulation circuit
To simulate the HS operation the commutation circuit
shown in Fig. 3 was modelled in PSpice software using
idealised switch models. The same diode model was used in
the topology for simplicity and the following simulation
parameters were assumed: the input voltage is 2800 V, the
maximum load current is 750. The values of the circuit’s
passive components are determined according to [9].
B. Control algorithm
Active states are generated using two-phase shifted triangle
waveform generators operating with constant frequency and
duty cycle and two comparators. Additional logic elements
ensure that the SB is turned on right at the instant the turn-off
of SA occurs (Fig. 5).
In real conditions the minimal and maximal duty cycle of the
HS could be limited by a number of factors, such as IGCT
gate driver limitations or behaviour of the clamp circuit and a
turn-off snubber (if used). The minimal off-state time toff(min)
should be maintained to stay within the safe operating area
(SOA) of the circuit’s components. The minimal on-state
time ton(min) of the HS is generally not limited since during an
operation with duty cycles near zero, only the transistor of the
HS could be used. The example of SOA control flowchart is
shown in Fig. 6.
Fig. 5. Control algorithm of the HS
Fig. 6. Generalised SOA control flowchart of HS
The simulations confirm the estimated behaviour of the
proposed switch configuration. At turn-on the HS operates
like an IGCT with the dI/dt clamp (Fig. 7). The on-state
voltage of the HS is equal to the voltage drop across the
thyristor during its conducting period (Fig. 8). During turn-
off of the HS the transistor is turned on for a short period; the
turning-off thyristor current is then transferred to the
transistor, which is closed right after the thyristor current
becomes zero. The turn-off dynamics of the HS are greatly
increased, while the excellent on-state characteristics of IGCT
remain (Fig. 9) and all the elements are operated within the
SOA.
IV. GENERALISED LOSS EVALUATION
The power loss estimation is one of the key points, crucial
for the circuit mechanical structure and the cooling system
design. The aim is to compare the power losses and
performance of the proposed switch configuration with
transistor- and thyristor-only counterparts at similar operation
parameters. In the comparison, a variable switched current of
350A < I < 900A and UDC=2800 V is assumed. The total
losses Ptot are calculated as the sum of conduction, turn-on
and turn-off losses at maximum junction temperature (125°C)
using the datasheet values of the devices.
0
500
1000
1500
2000
2500
3000
3500
1325 1327 1329 1331 1333 1335 1337 1339
Time (us)
U (V)
0
200
400
600
800
1000
1200
1400
1600
1800
I (A)
U (HS) I (SB ) I (SA)
0
1
1325 1327 1329 1331 1333 1335 1337 1339
SA(control)
I (SA)
U (HS)
Fig. 7. Simulated turn-on behaviour of HS at I=750 A, UDC=2800 V
0
1
2
3
4
5
6
7
8
9
10
390 400 410 420 430 440 450 460 470 480 490
Time (us)
U (V)
0
100
200
300
400
500
600
700
800
I (A)
U (HS) I (SB ) I (SA)
U (HS)
I (SA)
Fig. 8. Simulated conduction behaviour of HS at I=750 A, UDC=2800 V
362
In the simulations of losses, the minimum IGBT switching
losses with a very small gate resistances of RGon=4 Ω and
RGoff=2.5 Ω are assumed. In real industrial converters the
IGBT gate units are adjusted to generate the desired dI/dt and
dU/dt to avoid large voltage and current spikes during
transients. However, the use of the gate resistor to control the
dI/dt results in substantially higher switching losses in
IGBT [10]. If a dI/dt limiting turn-on snubber is used with
both IGBT and IGCT devices, the turn-on losses would be
similar [11]. On the other hand, the turn-off losses of the
device may increase slightly [12].
The turn-off losses of the IGCT were excluded in the
simulations; however, according to the test results presented
in the previous papers [13], the turn-off losses may not be
completely removed due to several factors. Firstly, for a large
area device, such as the IGCT, a significant output
capacitance must be charged in order to establish the
depletion region to support voltage. Another factor is the free
carriers which had not recombined being swept from the
junction. Nevertheless, an 89% reduction in turn-off losses
was reported in [14]. In real conditions, the power losses of
industrial applications could be distinctly higher than the
simulated values.
After the turn-on of the IGBT, the current distribution
between conducting IGBT and IGCT is mainly influenced by
different characteristics of the semiconductors, temperature
differences and asymmetrically distributed stray inductances
in the circuit [15][16] Assuming both semiconductors in
conducting state, the current sharing inside the HS neglecting
cell resistances and inductances can be calculated by
)(
)(
)(
IU
IU
I
I
k
T
satCE
SB
SA
I== (1)
0
1
480 490 500 510 520 530 540 550 560 570
SB(control)
0
1
480 490 500 510 520 530 540 550 560 570
SA(control)
0
500
1000
1500
2000
2500
3000
3500
4000
4500
480 490 500 510 520 530 540 550 560 570
Time (us)
U (V)
0
200
400
600
800
1000
1200
I (A)
U (HS) I (SB) I (SA )
I (SA) I (SB)
U (HS)
Fig. 9. Simulated turn-off behaviour of HS at I=750 A, UDC=2800 V
Using Eq. (1) the IGBT and IGCT currents could be
obtained by
I
SB k
II
+
⋅=
1
1 (2)
I
I
SA k
k
II
+
⋅=
1
(3)
According to simulations, the considered IGCT is showing
better dynamics for currents above 650 A, whereas the IGBT
is performing better at lower currents (Fig. 10). The proposed
switch configuration is estimated to provide 2.3…2.8 times
increased switching frequency in comparison to single hard
switched IGBT or IGCT with dI/dt clamp circuit exhibiting
the same power dissipation of 3 kW. Assuming the same
switching frequency in the range of 250…1050 Hz and
switch current of 750 A the IGBT performs better than IGCT
at frequencies above 450 Hz, whereas the HS provides
substantial (1.9…2 times) decrease in power losses in
comparison to single semiconductors (Fig. 11). Fig. 12 shows
average losses of all considered switch solutions operating in
the studied circuit with the wide range of duty cycles. The
IGBT performs better than IGCT up to D=0.85. Again, the
HS shows substantially (1.8…2.2 times) reduced power
dissipation in comparison to single semiconductors.
Unlike in the case of the typical parallel connection of
identical semiconductors, in the proposed HS both switches
are conducting full input current during the operation, thus
the current rating of both semiconductors must be sufficient.
On the other hand, the overall power dissipation is decreased
in comparison with single switches allowing one to increase
the switching frequency or reduce cooling system
requirements. Moreover, if one of the semiconductors fails,
the other one can still continue to operate independently
unless sufficient cooling is applied.
The economical feasibility of the HS implementation
greatly depends on the application and its operation
conditions. The comparison of semiconductor prices of
discussed switch configurations is shown in Fig. 13. It should
be mentioned, that semiconductor price is only a part of the
overall power electronic system. The prices of the passive
components greatly vary for different applications and are not
considered in this paper.
0
500
1000
1500
2000
2500
3000
350460570680790900
Switch Current (A)
Switching frequency (Hz)
IGBT IG CT HS
Fig. 10. Switch switching frequency vs. current for different semiconductor
configurations corresponding to 3 kW total power dissipation at
UDC=2800 V, D=0.5
363
0
1000
2000
3000
4000
5000
6000
250 350 450 550 650 750 8 50 950 1050
Switching freque ncy (Hz)
Average losses (W)
IGBT IGC T HS
Fig. 11. Switch power dissipation vs. switching frequency for different
semiconductor configurations at I=750 A, UDC=2800 V, D=0.5
1500
2500
3500
4500
5500
6500
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
Duty Cycle
Average losses (W)
IGBT IGC T HS
Fig. 12. Switch power dissipation vs. duty cycle for different semiconductor
configurations at I=750 A, UDC=2800 V, fsw=750 Hz
Fig. 13. Comparison of semiconductor prices of studied switch
configurations
V. CONCLUSION
Using commercially available 4.5 kV class IGBTs and IGCTs
in press-pack housings it is possible to create energy efficient
switches with essentially decreased power losses. Despite
having decreased maximum current capabilities in
comparison with parallel connected identical transistors or
thyristors and a higher price than single semiconductor
switches, the proposed switch configuration could be
beneficial in rolling stock converters or other applications
where higher switching frequencies are required or decreased
cooling system requirements are essential.
The future research will include construction of the hybrid
switch prototype, improvement of the simulation model
according to test results as well as investigation of the
benefits it could provide in modern converter topologies.
ACKNOWLEDGEMENT
This research work has been supported by Estonian
Ministry of Education and Research (Project SF0140016s11),
Estonian Science Foundation (Grants 7425, ETF8020) and
Estonian Archimedes Foundation (project „Doctoral school of
energy and geotechnology II“).
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