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Progress In Electromagnetics Research C, Vol. 18, 123–135, 2011
DESIGN OF A BALUN FOR A BOW TIE ANTENNA IN
RECONFIGURABLE GROUND PENETRATING RADAR
SYSTEMS
R. Persico
Istituto per il Beni Archeologici e Monumentali
Consiglio Nazionale delle Ricerche
Via Monteroni, Campus Universitario, Lecce 73100, Italy
N. Romano
Dipartimento di Ingegneria dell’Informazione
Seconda Universit`a degli studi di Napoli
Via Roma 29, Aversa 81031, Italy
F. Soldovieri
Istituto per il Rilevamento Elettromagnetico dell’Ambiente
Consiglio Nazionale delle Ricerche
Via Diocleziano 328, Napoli 80124, Italy
Abstract—This paper deals with the design of a reconfigurable
antenna that resembles a monolithic UWB bow-tie antenna for Ground
Penetrating Radar (GPR) applications. In particular, the attention
is focussed on the design of the balun system able to work in the
frequency band 0.3–1 GHz; the effectiveness of the design is shown by
examining the behaviour of the scattering parameters S11 for both the
reference monolithic antenna and the designed reconfigurable antenna.
Also, an analysis of the radiation pattern of both the monolithic and
reconfigurable antennas is presented and confirms the effectiveness of
the designed balun system.
Received 4 October 2010, Accepted 30 November 2010, Scheduled 9 December 2010
Corresponding author: Raffaele Persico (r.persico@ibam.cnr.it).
124 Persico, Romano, and Soldovieri
1. INTRODUCTION
Reconfigurable antennas are of great interest in communication
systems and in imaging and diagnostics, with reference both to single
antennas [1–3] and array of antennas [4, 5]. The flexibility of these
systems is, in general, exploited to change the work frequency band
and/or the direction of the main beam depending on the applicative
necessities.
Recently, the idea of the reconfiguration has been translated
also with regard to stepped frequency GPR systems with the aim of
achieving a large work band with relatively simple (linearly polarised
and electrically small) antennas. In this framework, some ideas and
also a first investigation of the relevant trade offs are reported in [6, 7];
in those papers, attention is devoted to the feeding path of the
antennas, which is switched vs. the frequency in order to improve
the matching of the antennas or at least to ensure a sufficient received
power on a large working frequency band.
With reference to the antenna to be designed in this work, which
is a bow-tie antenna, several interesting efforts have been made in
achieving improved performances about the UWB behaviour [8–10].
The possibility to make use of antennas with reconfigurable
radiating elements for GPR applications was presented in [11], where
the reconfiguration was performed by adopting the total geometry
morphing approach [12], The total geometry morphing approach [12]
represents the most structurally complicated but also the most versatile
method exploited to achieve antenna reconfigurability. Such an
approach is based on the interconnection of radiating elements very
small in terms of radiated wavelength that singly radiate in a non
efficient way. These sub-elements are interconnected by switches
(PIN diodes) in order to resemble the geometry of the antenna to
be synthesised. In this way, it is possible to change the antenna
shape in a controlled way vs. the frequency. In particular, in [11],
a bow-tie antenna was designed with the aid of the High Frequency
Simulator Software (HFSS), produced by Ansoft [13]. The total
geometric morphing allows to pursue the twofold goal of a constant
radiation pattern through the whole reconfigurable band and a good
matching at the same time, but of course presents also some drawbacks
(e.g., the effects of the non-ideal switch connections between the
“mosaic tiles” and the effects of the eddy currents excited on the
detached elements due to the incident field radiated from the connected
elements). For the designed reconfigurable bow-tie antenna, in [14, 15],
realistic operative condition, typical of the GPR applications, were
considered. In particular, in ref. [14], the designed antenna radiates
Progress In Electromagnetics Research C, Vol. 18, 2011 125
in presence of a receiving antenna within a configuration in reflection
mode, i.e., with the two antennas placed parallel to each other at the
interface of a half space.
All the previous papers were concerned with a simplified model of
the antenna where the problem of the feeding was not tackled. Here,
we deal with the problem of feeding the designed bow tie antenna from
a coaxial cable, and so the design of a balun system is needed too.
Therefore, the paper is organized as follow: In the next section,
the reference monolithic antenna and the discretized (reconfigurable)
version of it are briefly described. In Section 3 the exploited balun
system is presented. In Section 4, a comparison in terms of the
scattering parameter S11 (for different half-spaces) and radiation
pattern (in free space) for the monolithic and the discretized antennas
are shown. Conclusions follow in Section 5.
2. THE CONSIDERED RECONFIGURABLE ANTENNA
In this paper, we consider the reconfigurable antenna designed
in [11, 14] and we briefly report some details necessary for the correct
delivering of the paper.
The reconfigurable antenna design is based on the total geometry
morphing approach and for the case at hand a rectangular grid with
1295 pads and 2141 switches has been exploited. The size of the sub-
elements and interconnections has been designed to ensure a dynamic
structure easily adaptable to different geometries. In particular, each
sub-element is a square with side of 2.5 mm and is driven to the ON
or to the /OFF state by means of a ideal circuital model resembling
a typical PIN diode, whose resistance is R= 0.985 ∗10−3Ohm in the
ON state and R= 4 ∗106Ohm in the OFF state. PIN diodes sized
1×0.5 mm2have been exploited as interconnection elements.
The reconfigurable antenna was designed “to synthesise” a
reference monolithic bow-tie antenna that has a metallic element made
by copper with a thickness of 36 µm; the metallic element is located
on a layer of glass epoxy FR4 with extent 20 cm ×20 cm and 3 mm
thickness. The reconfigurable antenna in “bow-tie” configuration is
shown in Fig. 1(a); it has a flare angle of 30 degree. Fig. 1(b) depicts
a zoom of the feeding point for the reconfigurable antenna.
Figure 2 depicts the corresponding “monolithic” bow tie antenna,
whose details are reported in [11], for reference purpose with respect to
the comparison to be shown in the next sections. Let us outline that,
in principle, the physical differences between the two antennas are not
negligible, because the detached metallic elements of the discretized
bow tie enters in the field simulation too, as well as the fact that
126 Persico, Romano, and Soldovieri
the (many) present switches are not ideal but account for realistic
resistance values, as said. Finally, for simplicity, the small curvature of
the end of the arms of the monolithic antenna has not be translated into
the discretized reconfigurable structure, that ends with a flat bound.
3. RESULTS ABOUT THE DESIGNED BALUN
In this paper both the monolithic and the discretized antenna are fed
by means of a coaxial cable, as customarily happens in GPR systems
equipped with dipole-like antennas. The balancing of the currents
is achieved by means of the linearly tapered balun depicted in Fig. 3,
designed in planar technology [16] to achieve the balance of the currents
in the range 300–1000 MHz (leading to a fractional bandwidth equal
to 107.6%).
In particular, to ensure a gradual adjustment of the input
impedance of the antenna and the coaxial in the range 300 MHz–1GHz,
the length of the balun was chosen equal to L= 0.5λL, being λLthe
maximum work wavelength in the structure. This choice has been done
having in mind the design of an exponentially tapered transmission
line (which is a structure different but with some similarities with
the balun at hand). For a tapered transmission line connected to
a load impedance ZL, it is known that the impedance behaves as
an exponential function described by Z(z) = Z0eaz, in the range
0≤z≤L, where Z0indicates the characteristic impedance of the
line at z= 0 and a=1
Lln(ZL
Z0) [17]. It is intrinsically supposed that
the balun is matched with the impedance of the coaxial cable. The
(a) (b)
Figure 1. (a) Geometry of the reconfigurable antenna with pads
sized 2.5×2.5mm ×0.036 mm, interconnections by PIN diodes sized
1×0.5 mm and dielectric layer made with 16 ×19 cm ×3 mm glass-
epoxy FR4 Epoxy. The pads in “ON state” are highlighted in orange.
(b) Zoom of the feeding zone.
Progress In Electromagnetics Research C, Vol. 18, 2011 127
(a) (b)
Figure 2. (a) Reference bow-tie antenna: thickness 36 mm, radius
64 mm and flared angle 30 degree, placed on substrate epoxy FR4 with
extent 20 cm ×20 cm ×3 mm. (b) Zoom of the feeding zone.
reflection coefficient at distance Lfrom the load is given by [17].
Γ = ln ZL/Z0
2e−jβL sin βL
βL (1)
where β= (2π/λ) is the wave-number of the guided mode propagating
in the balun. Eq. (1) is based on the theory of the small reflections,
and so it is valid rigorously for small mismatching between load and
tapered line.
The modulus of the reflection coefficient in (1) is depicted in Fig. 4
and behaves as a sinc function; accordingly, if we increase the electrical
length of the taper so to guarantee βL ≥πin correspondence of the
minimum work frequency, this guarantees a low reflection all over the
work band, in the meaning that the reflection coefficient is attenuated
at least 13 dB compared to its value for a short length of the line. This
justifies the choice L= 0.5λL, where λLis the maximum wavelength
of the mode propagating along the tapered line.
In particular, we have made use of a substrate for the taper with
relative dielectric permittivity εr= 20 in the work frequency range
0.3–1 GHz, so to make shorter the wavelength within the taper; this
choice leads to a physical length Lequal to 26.8cm.
At this stage, however, we have still to match the taper to the
coaxial cable, i.e., we have to synthesise a value Z0(the characteristic
impedance of the line) close to the intrinsic impedance of the coaxial
cable, which is supposed here equal to 50 ohms. This aim is achieved by
designing suitably the widths of the arms of the taper. Let us note that
the proposed balun is similar to a micro-strip line on the unbalanced
side (see Fig. 3(a)) and to a double strip line at the gap of the bow-
tie antenna (see Fig. 3(b)). Therefore, the size of the microstrip line
on the unbalanced side (from the side of the coaxial cable) was set
128 Persico, Romano, and Soldovieri
(a)
(b) (c)
Figure 3. (a) Geometry of the linearly tapered balun. The double
stripline is connected to the antenna terminals; the microstrip line is
connected to the coaxial cable. (b) Detail of the connection coaxial
cable-balun. (c) Detail of the connection antenna-balun.
according to the following formula [14]:
W1
h=(8eA
e2A
−2per W1
h≤2
2
πnB−1−loge(2B−1) + εr−1
2εrCoper W1
h≥2)
where
A=Zc
60 rεr+ 1
2+εr−1
εr+ 1 µ0.23 + 0.11
εr¶
B=377π
2Zc√εr
C= loge(B−1) + 0.39 −0.61
εr
(2)
Progress In Electromagnetics Research C, Vol. 18, 2011 129
where W1is the width of the narrower conductor (the larger one does
not affect the impedance if it is sufficiently larger than W1), his the
transverse distance between the two conductors of the microstrip line
and Zcis the impedance of the coaxial cable (equal to 50 ohms as said
above). The distance hhas been set to 2.54 cm (it is the thickness
of the gap of the antenna), so W1has been chosen equal to 1.15 cm
according with Eq. (2). Finally W2has been chosen equal to 5W1thus
not affecting the validity of the formula in (2).
In order to test the designed balun, first the reflection coefficient
at the unbalanced end has been simulated, in the range 0.3–1 GHz.
The result is reported in Fig. 5 and shows that the matching with
the coaxial cable is very good all over the band 0.3–1 GHz and even
beyond.
Figure 6 shows instead the modulus of the reflection coefficient
at the end of the coaxial cable looking towards the antenna. More
precisely, the antenna is replaced by several test loads, in order to
check that the balun does not introduce (by itself) strong reflections
back into the coaxial cable. From this figure, it can be seen that the
balun does not introduce strong reflections by itself, even if it is not
able to match very well impedances strongly different from that of the
coaxial cable.
Of course Figs. 5 and 6 do not show the correct working of the
structure with regard to the balance of the currents between the
arms. In the next section, however, the comparison between the
radiation pattern of the monolithic and of the discretized structure
shall implicitly show the good quality of the balancing all over the
considered frequency band, inferred from the symmetric shape of the
radiation patterns.
1
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
0 5 10 15 20 25
Normalised units
β L
Figure 4. Modulus of the
reflection coefficient vs. βL
according to Eq. (1).
Figure 5. Frequency response
of the reflection coefficient at the
feed unbalanced port of the balun.
130 Persico, Romano, and Soldovieri
(a) (b)
(c) (d)
(e) (f)
Figure 6. Frequency response of the Modulus of S11 at the feed port
of the balun, when balun’s balanced load impedance is respectively:
(a) 50 Ohms. (b) 50 + j50 Ohms. (c) 75 Ohms. (d) 75 + j75 Ohms. (e)
200 Ohms. (f) 200 + j200 Ohms.
Progress In Electromagnetics Research C, Vol. 18, 2011 131
(a) (b)
(c) (d)
Figure 7. Frequency response of the Modulus of S11 at the feeded
port of balun (fed in unbalanced way by a coaxial cable). Blu line:
Monolithic antenna; Red line: Discretised reconfigurable antenna.
(a) Antenna in free-space. (b) Antenna on half space with relative
dielectric permittivity 4. (c) Antenna on half-space with relative
dielectric permittivity 9. (d) Antenna on half-space with relative
dielectric permittivity 16.
4. RESULTS ABOUT THE RECONFIGURABLE
ANTENNA
In this section some numerical results about both the discretized and
the monolithic antennas are shown, both with regard to the impedance
matching on a half space and the radiation pattern in free space. The
simulations have been performed making use of the HFSS code.
Figure 7 depicts the modulus of the reflection coefficient at the
input port of the balun, both for the discretized and the monolithic
antennas. As can be seen, the behaviour of the discretized antenna is
quite similar to that of the monolithic antenna. Noticeably, moreover,
132 Persico, Romano, and Soldovieri
the minima of all the reflection coefficients occurs about at the same
frequencies, which makes it possible to think of a relatively robust (with
respect to uncertainties about the permittivity of the soil) equalization
procedure.
Figure 8 shows the radiation pattern in free space of both the
monolithic and the discretised antennas. From Fig. 8, it can be
appreciated that the two patterns are in good agreement at 300
and 650 MHz. Some discrepancy arises at 1 GHz, where also some
asymmetric behaviour of the pattern (especially for the monolithic
antenna) can be seen. This is probably due to some current induced by
the field radiated by the arms on the external part of the coaxial cable.
In fact, the upper and lower lobe of the pattern on the E-plane remains
(a)
(b)
Progress In Electromagnetics Research C, Vol. 18, 2011 133
(c)
Figure 8. Radiation patterns in E-plane and H-plane normalized
to Emax in free space for the monolithic antenna (blue line) and the
discrete antenna (Red line). The feeding of the antenna comprises the
balun. (a) 300 MHz. (b) 650 MHz. (c) 1 GHz.
substantially equal to each other, and symmetrical with respect to the
270–90 degree axis, even if they are bent toward the direction of the
coaxial cable. The pattern at 300 and 650 MHz, instead, show the
typical shape of the pattern of a dipole like antennas. This is an
indirect proofs of the fact that the balance of the currents provided by
the balun is satisfying all over the band 300–1000MHz.
5. CONCLUSIONS
In this paper the discrete version of a bow tie antenna has been
proposed and compared with the corresponding monolithic antenna. It
has been shown that the discrete geometry provides results not much
different from the corresponding monolithic one. As a drawback, the
smallness of the elements of the discretised antenna makes it somehow
more complicated than the monolithic one, but it is also important
with regard to the mitigation of the spurious effects of the currents
excited on the detached elements of the discrete geometry.
This is a preparatory (and in progress) work in view of the
possibility to implement a reconfigurable antenna for GPR prospecting,
where the shape and the size of the discrete active elements can be
changed vs. the frequency in a controlled way by programming the
combined aperture of the switches. In this way one can extend the
134 Persico, Romano, and Soldovieri
equivalent work frequency band antenna both in terms of impedance
matching and in terms of radiation pattern. In fact, based on the
preliminary results shown here, we can expect that a suitable variation
vs. the frequency of the configuration of the active elements can
improve the matching meaningfully and can make the radiation pattern
less variable vs. the frequency.
As future research activity, we plan to implement the antenna
solution and perform realistic measurements to verify the effectiveness
of the proposed solution. For the realistic implementation of the
proposed solution it is necessary to design the DC network feeding
the PIN diodes and the cavity for the back lobe suppression.
ACKNOWLEDGMENT
The research leading to these results has received funding from the
European Community’s Seventh Framework Programme (FP7/2007-
2013) under Grant Agreement No. 225663 Joint Call FP7-ICT-SEC-
2007-1.
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