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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 70, NO. 12, DECEMBER 2021 12703
Wideband Gain Enhancement of an AMC
Cavity-Backed Dual-Polarized Antenna
Shuhui Yang, Member, IEEE, Longfei Liang, Wensong Wang , Member, IEEE, Zhongyuan Fang , Member, IEEE,
and Yuanjin Zheng , Senior Member, IEEE
Abstract—A novel low-profile wideband antenna is proposed
with high-gain, high-isolated, and dual-polarized properties. It con-
sists of two mutually-orthogonal bowtie-shaped dipole radiators,
four rectangular parasitic elements, and a cylindrical artificial
magnetic conductor (AMC) cavity. By introducing parasitic ele-
ments, additional resonance is generated, and the bandwidth is
widened. The gain is mainly improved by the cylindrical metallic
wall and wideband AMC of the cavity. The AMC operates in ±90o
reflection phase band-gap bandwidth of 2.35-4.25 GHz, and its
equivalent circuit model is developed to understand the working
mechanism. The orthogonal placement of two dipole radiators
ensures polarization diversity and high isolation. The impacts of
various environmental conditions on antenna performance are
investigated. To verify the concept of design and demonstrate its
excellent performance features, a prototype is fabricated and mea-
sured. The proposed wideband antenna has a height of 0.18 wave-
length at the center frequency. The measured -10-dB impedance
bandwidth is from 2.16 to 3.96 GHz, where the realized gain is
8.34-10.96 dBi and the radiation efficiency is more than 81.8%.
High isolation of >28.65 dB and low envelop correlation coeffi-
cient (ECC) of <2.74 ×10-3 are also achieved between antenna
elements. It can be used in the base station for intelligent Internet of
Vehicle (IoV) applications, covering the bands of WLAN, WiMAX,
Bluetooth, LTE, and 5G sub-6 GHz.
Index Terms—Dual-polarized high-isolated antenna, wideband
base station antenna, artificial magnetic conductor (AMC) cavity,
multiple input multiple output (MIMO), Internet of Vehicle (IoV).
I. INTRODUCTION
WITH the rapid development of fifth-generation (5G) and
beyond-5G wireless communication techniques, the fast
and reliable transmission rate can be enabled, which will signif-
icantly empower the Internet of Vehicle (IoV) applications [1].
The IoV is attractive as it will ensure the great convenience of
people’s daily activities. The coexisting of multiple wireless net-
works requires transceivers to work in multiple frequency bands
at the same time. As an essential part, the wideband antenna will
affect the wireless transmission signal’s speed and accuracy. The
Manuscript received August 15, 2021; revised October 6, 2021; accepted
October 10, 2021. Date of publication October 14, 2021; date of current version
December 17, 2021. The review of this article was coordinated by Dr. Xiaodai
Dong. (Corresponding author: Wensong Wang.)
Shuhui Yang and Longfei Liang are with the Department of Communication
Engineering, Communication University of China, Beijing 100024, China (e-
mail: yangshuhui@cuc.edu.cn; lianglf8305@163.com).
Wensong Wang, Zhongyuan Fang, and Yuanjin Zheng are with the
School of Electrical and Electronic Engineering, Nanyang Technologi-
cal University, Singapore 639798, Singapore (e-mail: wangws@ntu.edu.sg;
zfang005@e.ntu.edu.sg; yjzheng@ntu.edu.sg).
Digital Object Identifier 10.1109/TVT.2021.3119643
antenna needs to be specifically designed to enable the smooth
vehicle-to-vehicle/ vehicle-to-satellite communications, as well
as provide the wireless local area network (WLAN) in the vehicle
to meet drivers’ and passengers’ needs.
There are three main methods to achieve the wideband feature
for antennas. (i) The wideband feeding structure is utilized, such
as the Y-shaped feeding line [2], and the folded microstrip-line
balun integrated with a rectangular slot [3]. (ii) More resonance
modes are excited. By adding a shorting pin and V-shaped
slot to the equilateral triangular patch, the generated TM10,
TM20,TM
11 modes would result in the wideband property
[4]. Similarly, by introducing U-shaped slots to fan-shaped
radiators, the surface current path would be changed to produce
additional resonance, which reduces the radiation resistance to
be easily matched to 50 Ω; thus, the bandwidth enhancement
could be achieved [5]. (iii) Parasitic elements are placed with
optimal location and size, such as bowtie parasitic elements [6],
mushroom-like structures placed in the clearance area [7]. In [8],
the independent reflector and cross-π-shaped decoupling struc-
ture were added near the F-shaped radiator to generate two addi-
tional resonant frequencies, increasing the antenna bandwidth.
Also, high gain performance is an important indicator of
antenna designs. The antenna’s gain can be enhanced by em-
ploying parasitic/periodic structures, which are generally ar-
ranged above/under the radiator. In [9], the parasitic patch
was placed above the radiation direction, further concentrat-
ing the electromagnetic fields. In [10], the dual-sided partially
reflective surface (PRS) based on self-complementary struc-
ture was incorporated and located over with the radiator. Its
magnitude and phase can be tuned flexibly to raise the loaded
antenna gain. The nonuniform metasurface was further utilized
to design the broadband antenna [11]. Alternatively, the perfect
electric conductor (PEC) was often placed under the radiator
with a half-wavelength distance [2]. It reflects the backward
electromagnetic waves and makes them superimpose the direct
electromagnetic waves in front of the radiator. Instead of the
PEC ground plane, the artificial magnetic conductor (AMC) was
designed as a perfect magnetic conductor and it could reduce
the antenna profile [12], [13]. By using the AMC reflector, the
CP antenna were designed to achieve wider bandwidth [14]
and improve the gain [15]. Based on the AMC reflector, the
orthogonal differentially fed structure was designed to achieve
high isolation [16]. In [17], a dual-frequency AMC combined
with PIN diodes was introduced to realize a reconfigurable
antenna. As most of the vehicle-mounted antennas highlight
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12704 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 70, NO. 12, DECEMBER 2021
Fig. 1. Geometry of the proposed wideband antenna. (a) Top view. (b) Side
view. (c) AMC unit cell. (d) Vehicle-mounted antenna in intelligent IoV.
the feature of either wide bandwidth or high gain, it is highly
demanding and challenging to design a dual-polarized antenna
meeting both wide bandwidth and high gain, which can enable
the operation in multiple wireless communication bands of base
stations.
In this paper, a low-profile dual-polarized wideband an-
tenna is proposed for efficient wireless communication ap-
plications. Two bowtie-shaped dipole radiators are orthogo-
nally placed, which enhances the port isolation by provid-
ing polarization diversity. The rectangular parasitic elements
broaden the impedance bandwidth by creating additional reso-
nance. The cylindrical AMC cavity-backed structure, consist-
ing of a wideband AMC reflector and a cylindrical wall, is
employed to improve the loading radiator’s operating charac-
teristics in terms of profile reduction, gain enhancement, and
robustness to installation environment. Besides, the influence of
different environmental conditions on antenna performance is
studied. As a proof of concept, the proposed wideband antenna is
prototyped and measured. Referring to the presented guideline,
different types of antennas operating in other frequency bands
could be designed.
II. ANTENNA DESIGN
A. Configuration
Fig. 1(a) shows the structure of the proposed wideband an-
tenna. It consists of two mutually-orthogonal dipole radiators
(Antenna 1 and Antenna 2), four rectangular parasitic elements,
and a cylindrical AMC cavity. It is fed by using two coaxial
cables assembled with SMA connectors. The inner and outer
conductors of each coaxial cable are connected to two arms of
each dipole antenna, respectively. All copper patterns are printed
on the 1-mm thickness FR-4 substrate with a permittivity of 4.4,
loss tangent of 0.02. Four radiators are bowtie-shaped, each ro-
tationally distributed at 90°. Four rectangular parasitic elements
TAB LE I
GEOMETRIC PARAMETERS OF PROPOSED WIDEBAND ANTENNA (UNIT:MM)
are arranged in the middle between two radiators but locating
on the other surface of the substrate. Fig. 1(b) shows its side
view. The radiators are backed by a cavity, which is comprised
of a wideband AMC reflector and a cylindrical metallic wall. It
is noticed that a certain distance is kept between the radiators
and the AMC reflector. Fig. 1(c) shows the front view of the
wideband AMC unit cell consisting of three layers: an FR-4
dielectric layer for printing the metallic pattern, an air layer, and a
ground plate. The metallic pattern is comprised of four W-shaped
patches and a square ring. Geometric parameters of the proposed
wideband antenna are listed in Table I. Fig. 1(d) shows the
proposed wideband antenna’s potential application scenarios
for vehicular wireless communication. The vehicle-mounted an-
tenna functions as an important radiator for interacting with the
satellite and roadside unit (RSU). All simulations are conducted
by using the software ANSYS HFSS.
B. Evolution
The proposed wideband antenna is designed based on the
step-to-step evolution approach, which is shown in Fig. 2(a).
The four types of antennas have different heights. Type I is
comprised of Antenna 1 and Antenna 2, and both of them are
mutually-orthogonal. Type II is designed by diagonally arrang-
ing four rectangular parasitic elements between two adjacent
radiators of Type I, where the parasitic elements and radiators
are distributed on both sides of the substrate. Both antennas of
Type I and Type II are simulated without considering a ground
plane. A wideband AMC reflector is attached under the antenna
of Type II, which forms the antenna of Type III. In Type IV, a
cylindrical metallic wall is applied into the antenna of Type III.
The cylindrical metallic wall and the wideband AMC reflector
jointly contribute to the formation of a cylindrical AMC cavity.
The scattering parameters (S-parameters) are simulated for
different types of antennas, which are shown in Fig. 2(b). For
Type I, the impedance bandwidth of |S11|≤-10 dB is from 2.54
to 2.93 GHz, and a pair of dipole radiators generate the resonant
frequency f1at 2.71 GHz. For Type II, the impedance bandwidth
is from 2.26 to 3.84 GHz. f1moves to 2.51 GHz and another
resonant frequency f2is generated at 3.62 GHz by parasitic
elements. For Type III, the impedance bandwidth is from 2.19 to
3.78 GHz. f1and f2move to the lower frequency at 2.3 GHz and
3.35 GHz, respectively. For Type IV, the impedance bandwidth
is from 2.13 to 3.88 GHz. Besides f1and f2at 2.29 GHz and 3.8
GHz, respectively, the third resonant frequency f3is generated at
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YANG et al.: WIDEBAND GAIN ENHANCEMENT OF AN AMC CAVITY-BACKED DUAL-POLARIZED ANTENNA 12705
Fig. 2. (a) Evolution of the proposed wideband antenna. Simulated (b) S-
parameters, and (c) realized gain.
3.07 GHz. In all bandwidths, the port isolation |S21| is larger than
28.14 dB. As shown in Fig. 2(c), for Type I, the realized gain is
from 1.33 to 1.89 dBi in the bandwidth, and the maximum value
is at 2.8 GHz. For Type II, the parasitic elements improve the
realized gain to 2.16-3.04 dBi since the electromagnetic field
could be focused on the forward radiation, and the maximum
value is at 3.8 GHz. For Type III, the wideband AMC reflector
improves the realized gain to 6.94-9.32 dBi by reflecting the
backward electromagnetic field, and the maximum value is at
3.3 GHz. Moreover, the omni-radiation of Type II would become
the directional radiation of Type III. This operation is not only to
increase the gain, but also to reduce the impact from the vehicle
body. For Type IV, the cylindrical metallic wall improves the
realized gain to 7.62-11.54 dBi by reducing the side-lobe levels,
and the maximum value is still at 3.3 GHz. By using the designed
cylindrical AMC cavity, the realized gain is totally enhanced by
7.08-9.18 dBi.
III. PERFORMANCE ANALYSIS
A. Parasitic Elements
In Fig. 3, three parasitic elements in different shapes with
the same area are investigated for Type II: Triangular patches,
Trapezoidal patches, and Rectangular patches. When the par-
asitic element is triangular, the impedance bandwidth of |S11|
≤-10 dB is from 2.2 to 2.87 GHz, and the resonant frequency
is at 2.42 GHz. When the parasitic element is trapezoidal, the
impedance bandwidth is from 2.15 to 3.26 GHz, and the resonant
frequency is at 2.41 GHz. When the parasitic patch is rectangular,
the impedance bandwidth is from 2.23 to 3.8 GHz, and the
Fig. 3. Simulated |S11| for Type II with different parasitic elements.
Fig. 4. Simulated |S11| for Type II over (a) l5,and(b)l6.
Fig. 5. Simulated surface current distributions for Type II at (a) 2.56 GHz,
and (b) 3.75 GHz, where Antenna 1 is excited and Antenna 2 is terminated.
resonant frequencies can be observed obviously at 2.48 GHz
and 3.59 GHz. The structure can achieve a better bandwidth
and |S11|. The dimensions of rectangular parasitic elements are
further investigated, and the curves |S11 | over l5and l6are shown
in Fig. 4. As l5increases, |S11| becomes better; f1is almost
unchanged while f2moves to a low-frequency. As l6increases,
f1is kept unchanged, but f2will shift to a low-frequency, thereby
the bandwidth will become narrow. |S11| becomes larger at f1
while it becomes smaller at f2.
To better understand the operating principle of the parasitic
elements, the surface current distributions at 2.56 and 3.75 GHz
are illustrated in Fig. 5, where Antenna 1 is excited, and Antenna
2 is terminated to a load of 50 Ω. In Fig. 5(a), it is observed that
the surface currents mainly concentrate on the radiators of An-
tenna 1 at 2.56 GHz. At 3.75 GHz, the surface currents are more
dominant on both Antenna 1 and parasitic patches. They are
induced on the rectangular parasitic patches since the parasitic
elements are excited in a resonance state. The resonant frequency
is mainly related to the size of the parasitic elements. Therefore,
the lower frequency is generated by the driven bowtie-shaped
dipole radiators while the higher frequency is mainly generated
from the parasitic elements.
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12706 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 70, NO. 12, DECEMBER 2021
Fig. 6. (a) Evolution of the AMC pattern. (b) Simulated reflection phases.
B. Wideband AMC
To achieve a high-gain and low-profile antenna in a wide
bandwidth, the wideband AMC reflector is considered to
load below the radiator. As shown in Figs. 1(b) and (c), the
wideband AMC unit cell consists of metallic patterns, a metallic
ground plate, and the air layer between them. The metallic
pattern is printed on a 1-mm FR4 substrate, including a square
ring and four W-shaped patches surrounding it, which is kept at
a certain distance from the ground plate. The distance between
the wideband AMC reflector and the radiator is 8 mm (h1),
corresponding to 0.08 wavelength at 3.06 GHz.
The evolution of the metallic pattern is investigated, as shown
in Fig. 6(a). Case A includes a square ring, and four T-shaped
patches are attached to it. It is modeled with a Floquet Port
and a pair of Master-Slave boundary conditions for numerically
computing the reflection phase. Then, along two symmetry lines
perpendicular to each other, two rectangular slots are etched
on the pattern in Case A, and then the pattern becomes four
W-shaped sections, which is termed as Case B. By adding a
small square ring in the center, the pattern is further changed,
which is termed as Case C.
For the AMC unit cell, the in-phase reflection bandwidth, BW,
is defined as the reflection phase ranging from +90°to -90°.
The resonant frequency, f0, corresponds to the frequency at the
reflection zero-phase. The simulated BW and f0for three cases
are shown in Fig. 6(b). It is observed that they can move with
the change of the AMC pattern. For Case A, BW is from 1.91 to
2.95 GHz, and the fractional bandwidth (FBW) is 43.3%. Also,
f0is at 2.4 GHz. For Case B, BW is from 3.3 to 5.02 GHz (41.2%
FBW), and f0is at 4.17 GHz. Due to the addition of the slots,
the physical length is reduced by half, accordingly, f0moves to
a higher frequency, which is almost twice the original resonant
frequency. For Case C, BW is from 2.35 to 4.25 GHz (59%
FBW), and f0is at 3.22 GHz. Different from Case B, an inner
TAB LE I I
THE COMPARISON OF THE DESIGNED WIDEBAND AMC AND OTHERS
Fig. 7. (a) Equivalent circuit model, and (b) its corresponding component
values. (c) |S11|, and (d) reflection phase.
square ring is added in the center to increase mutual couplings
among W-shaped patches. This would reduce f0, accordingly,
the FBW becomes larger. As listed in Table II, a comparison of
the designed wideband AMC and other AMC structures is given
in terms of f0,BW, FBW, and unit cell patterns. It is observed that
the designed wideband AMC has a higher f0and larger FBW.
To further understand the wideband working mechanism of
Case C, the equivalent circuit model (ECM) is developed by
referring to the model presented in [18], as shown in Fig. 7(a).
With reference to several empirical formulas for calculating
lumped parameters [18], the initial values of these components
are assigned. The schematic circuit is simulated in the Advanced
Design System (ADS). After optimization, these corresponding
component values are summarized in the table in Fig. 7(b). In
Fig. 7(a), L1,C1, and R1are the inductance, capacitance, and
resistance for the square ring. L2,C2, and R2are the inductance,
capacitance, and resistance for the W-shaped patch. L4,C4, and
R4are the inductance, capacitance, and resistance between the
adjacent W-shaped patches. C3is the equivalent capacitance
between the adjacent AMC unit cells. L12,C12 , and R12 are the
inductance, capacitance, and resistance between the square loop
and W-shaped patches. Ldand Rdrepresent the inductance and
resistance from the FR4 substrate and the air layer, respectively.
Rgis the resistance of the copper ground. For the AMC unit cell,
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YANG et al.: WIDEBAND GAIN ENHANCEMENT OF AN AMC CAVITY-BACKED DUAL-POLARIZED ANTENNA 12707
Fig. 8. Simulated reflection phases over (a) s1,(b)s2,(c)s3,and(d)h2.
BW and f0could be calculated respectively by [19]
BW =1
η(Ld|| Lp)+L12
(CP+C12)(1)
f0=1
2π((Ld|| Lp)+L12)(Cp+C12 )(2)
where ηis the free space-wave impedance and it is 377 Ω.Cp
and Lprepresent the self-inductance and self-capacitance of the
metallic pattern, respectively, which could be expressed as
CP=C1+C2C4
C2+C4
+C3(3)
Lp=L1|| (L2+L4)(4)
The ADS circuit-based simulated |S11| and reflection phase are
shown in Figs. 7(c) and (d), respectively. Meantime, those results
derived from the HFSS model are also shown as a comparison.
It is found that the both results have a fairly good agreement
with each other. In Fig. 7(d), BW ranges from 2.3 to 4.3 GHz
in the HFSS model and it ranges from 2.35 to 4.36 GHz in the
ADS ECM. Both of the reflection zero-phases are at 3.2 GHz.
The values C3,C2,C1,L1,L2, and Ldare affected by s1,s2,s3,
and h2of the AMC, which are shown in Fig. 1, determining f0and
BW. Fig. 8 shows the simulated reflection phases of the wideband
AMC with varying s1,s2,s3, and h2. It is observed that the
reflection phase varies slightly with s1,s3, but violently with s2
and h2.Ass1or s2increases, f0moves to a high-frequency band,
and its corresponding BW also becomes wider. As s3increases,
f0moves to a low-frequency band, and BW becomes narrower.
As h2increases, f0moves to a low-frequency band, and BW first
becomes wider and then begins to be narrower. Referring to these
change trends of the structural parameters, BW and f0could be
adjusted to meet the design requirements. Through parametric
optimization, the reflection phase of +90°is at 2.3 GHz and the
one of -90°is at 4.4 GHz, and f0is located at 3.2 GHz.
Fig. 9. Simulated 3-D models and results with a sufficiently large reflector.
(a) The radiator is backed by the AMC reflector with the size of 616 mm ×616
mm. (b) The radiator is backed by the PEC reflector with the size of 616 mm ×
616 mm. (c) |S11| and (d) realized gain.
When the radiator is positioned close to the PEC reflector,
the radiation of the mirror currents on the PEC reflector will
cancel the radiation of the currents on the radiator since they
flow in opposite directions, and the radiation performance would
severely deteriorate. Thus, it requires sufficient spacing between
the radiator and the PEC reflector which is about quarter wave-
length [2]. Different from the PEC reflector, the AMC reflector
could produce the mirror currents with the same direction, and
thereby the forward radiation would be enhanced. Meantime,
the spacing between the radiator and the AMC reflector would
be reduced less than quarter wavelength. Fig. 9(a) shows that the
radiator is backed by the sufficiently large AMC reflector with
the size of 616 mm ×616 mm (6λ0×6λ0). By comparison,
the radiator is backed by the PEC reflector with the same size,
as shown in Fig. 9(c). When the similar |S11| and realized gain
are obtained as shown in Figs. 9(c) and (d), the height is 28 mm
(about λ0/4 at 3 GHz) for PEC while it is 18 mm for AMC.
By using the AMC reflector, the height of the antenna could be
reduced by 35.7%.
C. Cylindrical AMC Cavity
In the practical application, the size of the AMC reflector
should be comparable to the one of the radiator. When the size
of the reflector becomes small, the back-lobe and side-lobe levels
of the radiation would be large. By introducing the cylindrical
metallic wall to embrace the antenna with the AMC reflector, the
antenna’ realized gain is improved in the high-frequency band,
while the wide impedance bandwidth is unvaried, as shown in
Fig. 2. The cylindrical AMC cavity is formed by the cylindrical
metallic wall combined with the AMC reflector. It is helpful to
improve the performance of the antenna without increasing its
size.
Different number of AMC unit cells in the cavity is employed
to analyze the effect from 3 ×3, 4 ×4, 5 ×5, to 6 ×6, and
the simulated S-parameters results are shown in Fig. 10. It is
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12708 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 70, NO. 12, DECEMBER 2021
Fig. 10. Effect of the wideband AMC reflector in the cavity over unit cell
number. (a) |S11|, and (b) |S21|.
Fig. 11. Simulated 2-D radiation patterns of the antenna with/without cylin-
drical metallic wall at 3.1 GHz in (a) x-z plane and (b) y-z plane.
observed that there are three resonant frequencies in the range
of 1.5-5 GHz. When the unit cells vary from 3 ×3, 4 ×4, to
5×5, the second resonant frequency first moves to the low
frequency and then moves back to the high frequency. When the
unit cells vary from 5 ×5to6×6, the third resonant frequency
has a larger reflection coefficient. These phenomena are also
partly contributed by the radius change of the cylindrical metallic
wall. The port isolation is always larger than 25 dB over unit
cell number. Considering the AMC unit cells, wall radius, and
profile, a set of optimized AMC unit cells of 4 ×4 is selected
to obtain better antenna performance.
Keeping the wideband AMC reflector unchanged, the cylin-
drical metallic wall is removed. The simulated 2-D radiation
patterns of the antenna are shown in Fig. 11. Referring to them,
the proposed wideband antenna can obtain a better forward
radiation performance. The main-lobe level is improved by 1.56
dB at 3.1 GHz while the back-lobe level gets suppressed by
3.16 dB. Comparing with x-zplane, the y-zplane has a more
directive radiation pattern. In this design, the height (h3)ofthe
cylindrical metallic wall is optimized to stabilize the radiation
pattern. From Fig. 12, the presence of the cylindrical metallic
wall could concentrate the electric field distributions at the
forward radiation and reduce the back-lobe and side-lobe levels.
When the radius of the cavity is kept unchanged, four antennas
with different reflectors are discussed, including the antenna
with a PEC cavity of h3=18 mm, the antenna with a PEC
cavity of h3=27 mm, the antenna with an AMC cavity of
h3=18 mm, and the antenna without AMC/PEC cavity. The
former three kinds of antennas have similar gains while totally
different heights. Their simulated results are shown in Fig. 13.
When the antenna is with a PEC cavity, the height of the antenna
Fig. 12. Electric field distributions at 3.8 GHz for the antenna (a) without, and
(b) with cylindrical metallic wall. .
Fig. 13. Simulated results with no cavity, PEC cavity, and AMC cavity. (a)
|S11|, and (b) realized gain. .
would be 27 mm. By using the AMC cavity, the height of the
antenna (Type VI) is reduced to 18 mm, where a lower-profile
antenna could be obtained, and at the same time the realized
gain is almost unchanged in the same impedance bandwidth. By
using the PEC cavity of h3=18 mm, the bandwidth becomes
narrowest, and the realized gain is also reduced. After removing
the AMC/PEC cavity, the simulated |S11| and realized gain of
the antenna (Type II) are added for a comparison in Fig. 13. It
is observed that a narrower wideband and the lowest realized
gain are obtained. Therefore, the cylindrical AMC cavity could
contribute to the improvement of the bandwidth and gain of the
antenna.
IV. IMPACT OF ENVIRONMENTAL CONDITIONS ON ANTENNA
PERFORMANCE
For the practical working environment, the antenna perfor-
mance would be affected by the surface of the truck roof, which
could be approximately regarded as a large metallic plate. When
the antenna is mounted on the vehicle, the surface currents would
be induced on the conductive objects. Moreover, the transmitting
and receiving circuits would occupy some spaces between the
antenna and installation location. A suitable spacing from the
metallic surface of the vehicle is required to keep the optimal
antenna performance. To reduce the computational volume, the
surface of the truck roof is imitated by a square metallic plate
with the width (W) of 450 mm and thickness of 1 mm, as shown
in Fig. 14(a). The proposed wideband antenna is placed over the
metallic plate, and the distance (H) varies from 0, 10, 20, to 30
mm. The coaxial cables with SubMiniature-version-A (SMA)
connectors are used as the feedings but they are not grounded to
the large metallic plate, as shown in Fig. 14(b). The simulated
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YANG et al.: WIDEBAND GAIN ENHANCEMENT OF AN AMC CAVITY-BACKED DUAL-POLARIZED ANTENNA 12709
Fig. 14. The proposed wideband antenna and imitated vehicle metallic surface.
(a) Simulated model, and (b) side view.
Fig. 15. Simulated reflection coefficients and port isolation over H.(a)|S11 |
and (b) |S21|.
Fig. 16. Simulated 2-D radiation patterns over Hat 3.1 GHz. (a) x-z plane and
(b) y-z plane.
reflection coefficients and port isolation are shown in Fig. 15.
With the increase of H,the-10-dB impedance bandwidth and
port isolation are almost unchanged. The influence could be
ignored to the distance change between the antenna and metallic
plate. The simulated 2-D radiation patterns of the antenna with
different distances at 3.1 GHz are shown in Fig. 16. It is observed
that they could be kept stable. When H=20 mm, and Wvaries
from 250, 350, 450, to 550 mm, the simulated reflections coeffi-
cients and port isolation are shown in Fig. 17. The simulated 2-D
radiation patterns of the antenna with different distances at 3.1
GHz are shown in Fig. 18. By comparisons, the antenna perfor-
mance remains almost unchanged. Stable antenna performance
is robust to the installation location. This will greatly reduce the
installation time and location selection.
When Antenna 1 is excited by 1-W source power, the surface
current density distribution is shown in Fig. 19(a), where it
mainly distributes along the x-axis, and the maximum density
is 8.9127 ×10-2 A/m. In the simulations, the feeding ports are
not grounded to the large metallic plate and the setup can refer
to the presentation in Fig. 14(b), where H=20 mm. When
Fig. 17. Simulated reflection coefficients and port isolation over W.(a)|S11|
and (b) |S21|.
Fig. 18. Simulated 2-D radiation patterns over Wat 3.1 GHz. (a) x-z plane and
(b) y-z plane.
Fig. 19. Surface current density distributions at 3 GHz. (a) Antenna 1 is
excited, (b) Antenna 2 is excited, and (c) both are excited for the proposed
wideband antenna. (d) Antenna 1 is excited, (e) Antenna 2 is excited, (f) both
are excited for Type II.
Antenna 2 is excited by 1-W source power, the surface current
distribution is shown in Fig. 19(b). It is observed that it mainly
distributes along the y-axis and the maximum density is 9.372
×10-2 A/m. When Antenna 1 and Antenna 2 are excited by
1-W source power, the surface current distribution is shown
in Fig. 19(c). It is noticed that it mainly distributes along the
included angle between the x-axis and y-axis, and the maximum
density is 1.2573 ×10-1 A/m. When the cylindrical AMC cavity
is removed, the antenna structure becomes Type II. It is placed
at 18 mm away from the metallic plate. When Antenna 1 is
excited by 1-W source power, the surface current distribution is
shown in Fig. 19(d). It mainly distributes along the y-axis, and
the maximum density is 7.2849 ×10-1 A/m. When Antenna 2 is
excited by 1-W source power, the surface current distribution is
shown in Fig. 19(e). It is noticed that it mainly distributes along
the x-axis and the maximum density is 7.9286 ×10-1 A/m. When
Antenna 1 and Antenna 2 are excited by 1-W source power, the
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12710 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 70, NO. 12, DECEMBER 2021
TABLE III
DIFFERENT PARAMETERS OF SURROUNDINGS
Fig. 20. 3-D radiation patterns of the proposed wideband antenna placed on
the truck roof in different surroundings. (a) Normal air, (b) dry air, and (c) humid
air at 2.4 GHz. (d) Normal air, (e) dry air, and (f) humid air at 3.5 GHz.
surface current distribution is shown in Fig. 19(f). It is seen that
it mainly distributes along the included angle between the x-axis
and y-axis, and the maximum density is 1.0386 A/m.
Different environmental conditions are imitated by setting
different parameters of surroundings in the simulation software,
and these specific parameters are listed in Table III. The proposed
wideband antenna is placed on the truck roof. The vehicle-
mounted antenna could work in different environmental con-
ditions such as normal air, dry air, and humid air environments.
The 3-D radiation patterns of the proposed wideband antenna
at 2.4 and 3.5 GHz are shown in Fig. 20. Due to rotationally
symmetrical geometry of the antenna, only Antenna 1 is excited,
and the other is terminated with the load of 50 Ω. In the normal
air, the gain is 9.7909 dBi at 2.4 GHz and 11.5683 dBi at 3.5
GHz. When there is no truck, the simulated gain of the proposed
wideband antenna is 9.1554 dBi at 2.4 GHz and 10.98057 dBi
at 3.5 GHz. In the dry air, the gain is 9.7913 dBi at 2.4 GHz and
11.5652 dBi at 3.5 GHz. In the humid air, the gain is 9.7297 dBi at
2.4 GHz and 11.5207 dBi at 3.5 GHz. For different environmen-
tal conditions, the gain of the antenna changes slightly, but the
pattern has no obvious change. A good unidirectional radiation
is achieved, and the vehicle-mounted antenna can maintain good
radiation characteristics. The proposed wideband antenna could
be easily mounted on the metal surface without affecting the
high isolation, wide bandwidth, and high gain.
V. M EASUREMENT AND COMPARISON
The prototype of the proposed wideband antenna is fab-
ricated, and its front view and side view are shown in the
Fig. 21. Fabricated prototype (Right insets: front view and side view) and the
measurement in the anechoic chamber.
Fig. 22. Measured and simulated S-parameters.
Fig. 23. Measured and simulated (a) realized gain and radiation efficiency,
and (b) ECC for the proposed wideband antenna.
right insets in Fig. 21. Four plastic posts are used to support
different substrates. The height of the cylindrical AMC cavity
is 0.18λ0(λ0is the free-space wavelength at the center fre-
quency). The measurement in an anechoic chamber is shown in
Fig. 21. The S-parameters are measured by an Agilent E5071C
vector network analyzer. As shown in Fig. 22, the measured
impedance bandwidth of |S11|≤-10dBisfrom2.16to3.99GHz,
and the simulated result is from 2.14 to 3.92 GHz. The measured
impedance of |S22|≤-10dBisfrom2.15to3.96GHz,and
the simulated result is from 2.13 to 3.89 GHz. Overlapping
measured bandwidth is from 2.16 to 3.96 GHz. There are three
tips at around 2.34, 2.93, 3.86 GHz. The measured isolation |S12|
is more than 28.65 dB, and the simulated result is more than
28.58 dB. The slight discrepancy between the simulated and
measured results is caused by the cable loss, SMA connector,
and welding process. Fig. 23(a) presents the measured and sim-
ulated realized gain and the radiation efficiency of the proposed
wideband antenna. The measured results and simulated data are
consistent overall. The measured gain is from 8.34 to 10.96 dBi,
and the maximum value is at 3.4 GHz. The simulated gain is from
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YANG et al.: WIDEBAND GAIN ENHANCEMENT OF AN AMC CAVITY-BACKED DUAL-POLARIZED ANTENNA 12711
Fig. 24. Measured and simulated 2-D radiation patterns in the x-z plane and
y-z plane at (a) 2.4 GHz, and (b) 3.5 GHz.
TAB LE I V
THE COMPARISON OF THE PROPOSED WIDEBAND ANTENNA AND OTHERS
8.48 to 11.49 dBi, and the maximum value is at 3.3 GHz. The
measured efficiency is 81.8-95.4%, and the simulated efficiency
is 82.04-96.48%.
The envelope correlation coefficient (ECC) is employed to
evaluate the degree of independence between antenna elements,
which could be computed by using the S-parameters, as follows
ECC =|S∗
11S12 +S∗
21S22 |2
1−|S11 |2−|S21|21−|S22|2−|S12|2(5)
As shown in Fig. 23(b), the measured ECC is from 2.26
×10-5 to 2.74 ×10-3, and the simulated ECC is from 2.17
×10-6 to 2 ×10-3. It means that Antenna 1 and Antenna
2 can work independently. Fig. 24 shows the measured and
simulated co-polarized and cross-polarized radiation patterns in
the x-z plane and y-z plane at 2.4 and 3.5 GHz. The radiation
patterns are stable and generally symmetrical with the main
beam fixed at boresight. The cross-polarizations are all less than
35 dB below the co-polarization level. As listed in Table IV, a
comparison between the proposed wideband antenna and other
similar antennas is given in terms of center frequency (CF),
FBW, height, realized gain, port isolation, maximum aperture
efficiency (MAE), and design approach. The proposed wideband
antenna has a wider bandwidth, higher realized gain, and larger
MAE. Its overall height has been reduced due to the use of
the designed wideband AMC. With these good features, the
proposed wideband antenna can be used in base stations for
IoV, covering the bands of WLAN (2.4-2.4835 GHz), WiMAX
(2.1-3.5 GHz), Bluetooth, LTE (2.2-3.8 GHz), and sub-6 GHz
of 5G (3.3-3.6 GHz).
VI. CONCLUSION
In this paper, a low-profile dual-polarized antenna has been
presented with wide bandwidth and high gain. The wideband
mechanism and high-gain characteristics are studied, including
both theoretical analysis and experimental verification. The
bowtie-shaped radiator generates the first resonant frequency.
By loading with rectangle parasitic elements, the additional
resonant frequency is created. Two resonant frequencies jointly
contribute to the wideband characteristic. The cylindrical AMC
cavity-backed structure improves the reflection coefficient and
the realized gain. The impacts of different environmental con-
ditions on antenna performance are analyzed. The fabricated
prototype has been measured, and the measured results agree
well with the simulated data. The proposed wideband antenna
can cover the band from 2.16 to 3.96 GHz, FBW 58.82%. Its
excellent performance is achieved with the realized gain of
8.34-10.96 dBi, the radiation efficiency of 81.8-95.4%, the ECC
of less than 2.74 ×10-3, and the port isolation of higher than
28.65 dB within the operating frequency band. These excellent
features demonstrate that the proposed wideband antenna can
be used in multiple wireless communication bands for IoV.
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Shuhui Yang (Member, IEEE) received the B.Sc.
degree from Zhejiang University, Hangzhou, China,
in 1994, the M.Sc. degree from the Communication
University of China, Beijing, China, in 2000, and the
Ph.D. degree from the Institute of Microelectronics,
Chinese Academy of Sciences, Beijing, China, in
2003. From 2003 to 2015, he was with Beijing In-
formation and Science Technology University, Bei-
jing, China, as a Professor. In 2015, he joined the
Communication University of China, and is currently
a Professor and the Chairman of the Department of
Communication Engineering. In 2008, he was a Visiting Scholar and an Adjunct
Associate Professor with the University of South Carolina (USC), Columbia, SC,
USA. In 2011, he was a Visiting Scholar with Victoria University, Melbourne,
VIC, Australia. From 2013 to 2014, he was a Visiting Professor with USC. His
research interests include electromagnetic metamaterials, passive microwave
components, electromagnetic compatibility (EMC), signal integrity (SI), and
wireless inter-/intra- chip communication system. He is a Senior Member of the
Chinese Institute of Electronics (CIE).
Longfei Liang received the B.Sc. degree in Internet
of Things engineering from Yantai University,Yantai,
China, in 2019. He is currently working toward the
M.Sc. degree with the Communication University of
China, Beijing, China. His research interests include
artificial magnetic conductor and wideband MIMO
antenna.
Wensong Wang (Member, IEEE) received the Ph.D.
degree from the Nanjing University of Aeronautics
and Astronautics, Nanjing, China, in 2016. From
2013 to 2015, he was a Visiting Scholar with the
University of South Carolina, Columbia, SC, USA.
In 2017, he joined Nanyang Technological Univer-
sity, Singapore, as a Research Fellow, and he is cur-
rently a Senior Research Fellow. His research inter-
ests include RF/microwave components and systems,
inter/intra-chip wireless interconnect, power wireless
transfer, signal integrity, and non-destructive real-
time health monitoring.
Zhongyuan Fang (Member, IEEE) received the
B.Sc. degree in microelectronics from Fudan Uni-
versity, Shanghai, China, in 2016 and the Ph.D.
degree from Nanyang Technological University,
Singapore, in 2021. His research interests in-
clude analog/mixed-signal integrated circuit design,
energy-efficient algorithms for physiological signal
processing, low-power sensor interface IC design,
and also RF/Wireless communication circuits and
systems design and testing for portable biomedical
applications.
Yuanjin Zheng (Senior Member, IEEE) received the
B.Eng. and M.Eng. degrees from Xi’an Jiaotong Uni-
versity, Xi’an, China, in 1993 and 1996, respectively,
and the Ph.D. degree from Nanyang Technological
University, Singapore, in 2001. From July 1996 to
April 1998, he was with the National Key Labora-
tory of Optical Communication Technology, Univer-
sity of Electronic Science and Technology of China,
Chengdu, China. In 2001, he joined the Institute of
Microelectronics (IME), Agency for Science, Tech-
nology and Research (A∗STAR), and is a Principle
Investigator and Group Leader. In July 2009, he joined Nanyang Technological
University, and is currently an Associate Professor and the Director of the Center
of Integrated Circuits and Systems. He has authored or coauthored more than
400 international journal and conference papers, 26 patents filed, and five book
chapters. His research interests include integrated circuits design, 3D imaging
and display, and SAW/BAW/MEMS sensors for NDT. He has led and completed
projects by working with industry partners and developing commercial products.
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