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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 69, NO. 11, NOVEMBER 2021 7323
Compact Dual-Polarized Wideband Antenna With
Dual-/Single-Band Shifting for Microbase
Station Applications
Wensong Wang ,Member, IEEE, Shuhui Yang, Member, IEEE, Zhongyuan Fang ,Member, IEEE, Quqin Sun,
Yinchao Chen, Senior Member, IEEE, and Yuanjin Zheng ,Senior Member, IEEE
Abstract— A gain-enhanced wideband multiple-input multiple-
output (MIMO) antenna with dual polarization and high isolation
is proposed for microbase station application. It includes two
mutually orthogonal bowtie dipole radiators, four bowtie par-
asitic elements with the switch function of ON-/OFF-state, four
vertical stubs, and a cavity, which are fed by a pair of 50
coaxial cables. Shifting between the dual-band and single-band
is realized by the switch. Four resonant frequencies are generated
by cooperative operation of the radiators and cavity, and thus
two wide bands are formed. C-shaped slots on radiators and
switch slots on parasitic elements are analyzed for estimating the
impact on resonant frequency and bandwidth, and the vertical
stubs improve the impedance matching. The proposed wideband
antenna achieves enhanced diversity performance in the MIMO
antenna system. As a proof of concept, a prototype of the
expanded four-element MIMO antenna is fabricated. Measured
results match well with simulated data. When the switch is
ON-state, it operates in 1.58–2.77 and 4.71–6.18 GHz with gains
of 9.4–11.2 and 9.1–10.9 dBi, and radiation efficiencies of ≥87.5%
and ≥87%, respectively. When the switch is OFF-state, it operates
in 1.56–2.76 GHz with a gain of 8.9–10.1 dBi and radiation effi-
ciency of ≥88%. Element isolation is ≥35.3 dB, and envelope cor-
relation coefficient (ECC) is ≤0.002. The experiment of the estab-
lished 2×2 MIMO system demonstrates that it can operate with a
lower bit error rate (BER), and the design shows enormous poten-
tial to be applied as massive MIMO microbase station antennas.
Index Terms—Cavity-backed antenna, dual-/single-band shift-
ing, high-isolated and dual-polarized antenna, Internet of Vehi-
cles (IoV), microbase station, multiple-input multiple-output
(MIMO).
I. INTRODUCTION
LARGER wireless data transfer and more flexible adapt-
ability to the exponential traffic environment are currently
in great demand. As multiple-input multiple-output (MIMO)
Manuscript received February 9, 2021; accepted April 8, 2021. Date
of publication May 4, 2021; date of current version October 28, 2021.
(Wensong Wang and Shuhui Yang contributed equally to this work.)
(Corresponding author: Wensong Wang.)
Wensong Wang, Zhongyuan Fang, Quqin Sun, and Yuanjin Zheng
are with the School of Electrical and Electronic Engineering, Nanyang
Technological University, Singapore 639798 (e-mail: wangws@ntu.edu.sg;
zfang005@e.ntu.edu.sg; ernestsun@ntu.edu.sg; yjzheng@ntu.edu.sg).
Shuhui Yang is with the Department of Communication Engineering, Com-
munication University of China, Beijing 100024, China (e-mail: yangshuhui@
cuc.edu.cn).
Yinchao Chen is with the Department of Electrical Engineering, University
of South Carolina, Columbia, SC 29208 USA (e-mail: chenyin@cec.sc.edu).
Color versions of one or more figures in this article are available at
https://doi.org/10.1109/TAP.2021.3076256.
Digital Object Identifier 10.1109/TAP.2021.3076256
system can improve the spectrum usage efficiency, commu-
nication quality, data rate, and reliability, MIMO antennas
have been widely studied to serve as an important device for
microbase stations in Internet of Vehicles (IoV) [1]. By lever-
aging the same frequency and time resources, MIMO antennas
can transmit and receive the electromagnetic wave energies via
complex scattering propagation environments. Considering the
coexistence of multiple wireless networks, it is preferred that
these antennas can work in different bands. Compared with
the integrated circuits for data processing and analysis, these
antennas are still occupying a larger area of the system [2].
When the space of deploying multiple antennas is reduced,
mutual coupling poses a severe challenge [3]. In practice,
intelligent edge devices do not need to communicate with
microbase stations all the time; thus, they could keep in touch
at a lower data rate and security [4], [5]. Once needing a com-
munication with higher data rate and security, an alternative
frequency band is timely enabled to ensure high-efficiency data
transmissions. Moreover, the dual-band feature is promising
to suppress the effects of interferences in communication
systems, where one of them is used in normal communications
and the other can switch in two modes with the ON-state
and OFF-state according to the requirements of data rate and
security level.
Dual-band MIMO antennas are attracting the attention of
research communities considering the structural simplicity and
easy implementation. Dual-band features can be achieved in
three approaches. One is to apply multiple radiators, which
create multiple resonances such as dual-band planar inverted-F
antennas [6]–[8]. Length-different slots were employed in the
microstrip-line-fed antenna, which works at 2.5 and 5.6 GHz
[9]. Two different bowtie dipoles were arranged back to back,
which forms two bands [10]. However, the occupied area
increases rapidly with an increase of structural complexity
for most of these designs. The second approach is to utilize
the internal coupling effect. A parasitic rectangular patch
was coupled with the meander-line antenna to achieve dual-
band operation [11]. By using the edge-to-edge coupling
between folded Y-shaped isolator and the meandered mono-
pole radiators, an extra lower-band resonant frequency was
achieved [12]. The third approach is to employ multiple
modes generated from the perturbation patch/slot or high-
order mode. After the L-shaped slot was engraved in the
radiating patch, the second harmonic frequency was lowered
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7324 IEE E TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 69, NO. 11, NOVEMBER 2021
to the band of interest [13]. By introducing a via to modify
the resonant mode, the substrate-integrated-waveguide-based
textile antenna could work in two wide bands [14]. The last
approach is widely used since the design procedure can be
found for quantitative analysis.
When element isolation of around 20 dB was obtained in
free space, the physical separation should be larger than half-
wavelength [15], which is not feasible for high-integration
space-constrained MIMO systems. There are two methods
utilized to mitigate undesired mutual coupling and thus raise
interelement isolation levels. One method is that an appropriate
grounded pattern is placed between antenna elements by
inserting slots [9], metalized walls [6], [13], electromagnetic
bandgap (EBG) [11], and other shapes of isolator structure
[7], [8], [12]. The other is that the interelement orthogonal
arrangement is considered for MIMO antennas where the
surface current directions on two radiators are perpendicular
to each other [10], [14], [16].
By analyzing dual-band formation and decoupling method,
the synergistic combination of orthogonal arrangement and
higher harmonics is considered, which achieves high isolation
and compact size. In this article, an innovative wideband
MIMO antenna is proposed for microbase station applications,
which can shift in the dual-band and single-band with dual
polarization, high isolation, and high gain. The working mech-
anism of resonant frequency generation and wideband forma-
tion is investigated, and the effects of C-shaped slots, switch
slots, and switch placement are studied. Diversity performance
analysis is conducted to discuss the performance of the pro-
posed wideband antenna in the MIMO system. As a proof
of concept, the expanded four-element array of the proposed
wideband antenna is fabricated, and the measurements are
carried out in the anechoic chamber. Attractive performances
are highlighted by a comparison. At last, the 2 ×2MIMO
system is established for further verifying the practicality of
the antenna.
II. DESIGN CONCEPT AND ANTENNA CONFIGURATION
A. Wideband Antenna Configuration
As shown in Fig. 1(a), the proposed wideband antenna is
comprised of a pair of mutually orthogonal bowtie dipole
radiators (Antenna 1 and Antenna 2), four bowtie parasitic
elements with switches, four vertical stubs, and a cavity. The
spacing between the two elements is zero due to orthogonal
arrangement. It is feasible to produce a lower mutual cou-
pling for high-integration space-constrained MIMO terminals.
Efficient isolation enhancement can be achieved. Two coaxial
lines with characteristic impedances of 50 are employed to
connect the feeds of these radiators. The C-shaped slots are
etched on the radiating elements, and each radiating element
is connected to a vertical stub in parallel at the corresponding
connected port. The switch slots are etched on the parasitic
elements. Adjacent radiating element and parasitic element
are spatially distributed at 45◦and, respectively, printed on
the top and bottom surfaces of a 1 mm-thick FR-4 epoxy
substrate with the relative permittivity of 4.4 and loss tangent
of 0.02. Four vertical stubs are also printed on FR-4 epoxy
Fig. 1. Geometry and dimensions of the proposed wideband antenna.
(a) Overall view. (b) Zoomed-in view between Antennas 1/2 and coaxial cables
1/2. (c) Front view. (d) Vertical stub where a=16, b=4, d=5.2, e=18,
g=1, h=5.5, k=4, l=13.5, m=14, n=1.2, p=1.3, q=23.5,
o=6, R=53, S=12.7, u=1.4, v=1.6, and H=40, all in mm.
Fig. 2. Parasitic element with the switch. (a) Switch slot. (b) ON-state
equivalent model. (c) OFF-state equivalent model.
substrates, but the thickness is 0.4 mm. As shown in Fig. 1(b),
one of the bowtie radiating patches of each antenna element
is connected to the inner conductor of the semirigid coaxial
cable, while the other is connected to the outer conductor.
Fig. 1(c) and (d) shows the detailed structural parameters of
the proposed wideband antenna, whose overall size is with a
diameter of 106 mm and a height of 40 mm. As illustrated in
Fig. 2(a), each parasitic element with the open-ended straight
slot and a switch can be equivalent to two kinds of models.
When the switch is in the ON-state, the slot on the parasitic
element is disconnected, termed a shorted slot, which is
illustrated in Fig. 2(b). When the switch is in the OFF-state,
the slot on the parasitic element is connected, termed an open
slot, which is illustrated in Fig. 2(c).
B. Resonant Frequency Generation
To further analyze the switchable wideband characteristic
of the proposed wideband antenna, the working mechanism
is investigated. A pair of mutually orthogonal dipole radi-
ators with different surroundings are modeled, as shown in
Fig. 3. Case 1 consists of two mutually orthogonal dipole
radiators (Antenna 1 and Antenna 2), which are bowtie-
shaped with each distributed at 90◦rotationally. Case 2 is
designed by enclosing these radiators into a cylindrical electric
wall. The height of the wall is about a quarter free-space
wavelength at the resonant frequency, and the radius depends
on the dimension of the radiators. Case 3 is designed by
placing a circular metallic plate under these radiators with
a distance of a quarter free-space wavelength at the resonant
frequency. Case 4 is achieved by using a cavity composed
of the cylindrical electric wall and the circular metallic plate.
Simulated reflection coefficients of Cases 1, 2, 3, and 4 are
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WANG et al.: COMPACT DUAL-POLARIZED WIDEBAND ANTENNA WITH DUAL-/SINGLE-BAND SHIFTING 7325
Fig. 3. Pair of mutually orthogonal dipole radiators with different surround-
ings. (a) Case 1. (b) Case 2. (c) Case 3. (d) Case 4.
Fig. 4. Refection coefficients of a pair of mutually orthogonal dipole radiators
with different surroundings.
shown in Fig. 4. It can be found that Case 1 gives two
resonant frequencies of 2.15 and 6.75 GHz, recorded as f1and
f2, respectively. The design procedure of the bowtie-shaped
radiator is similar to the design of a rectangular microstrip
antenna, the resonant frequency f1can be obtained using the
equations [17], [18]
f1=c
2√εeL·1.152
Rt
(1)
Rt=L
2·(a+2L)+(b+2L)
(a+2L)(2e+2L)(2)
L=t·0.412(εe+0.3)Wi
t+0.262
(εe−0.258)Wi
t+0.813(3)
εe=εr+1
2+(εr−1
2)(1+12t
Wi
)−1/2
(4)
Wi=a+b
2(5)
where t,εr,andεeare the thickness, relative, and effective
permittivity of the substrate, respectively. In Fig. 4, it is found
that the first band is from 2 to 2.3 GHz with a fractional
bandwidth (FBW) of 13.95% for Case 1. Thus, a common
bowtie-shaped antenna may not be able to provide a wide
bandwidth, which is less than 20% FBW. f2is the second har-
monic frequency which can be expressed as f2=αf1where α
is the frequency ratio factor, herein it is 3.2 [19]. The second
band is from 6.55 to 6.95 GHz, FBW of 5.9%. Case 2 indicates
that besides f1of 2.25 GHz and f2of 6.55 GHz, the third
resonant frequency f3is created at 5.15 GHz since the hollow
cavity surrounded by a cylindrical-shaped metal conductor can
confine electromagnetic waves by reflecting them back and
forth between the cavity’s boundaries. Currents in the walls
build up standing electromagnetic waves that form specific
resonant modes at certain frequencies, and thereby, the cavity
stores electromagnetic energy. For a cylindrical cavity of
radius Rand height H, the frequency of resonance modes
is given as
fTM/TE
mnl =c
2π√μrεr(p
mn
R)
2
+lπ
H2
(6)
where εrand μrare the relative permittivity and permeability,
respectively, p
mn is the mth root of the Bessel function
J
nof the first kind, and m,n,andlrefer to the number
of half-wavelength variations in the standing-wave patterns
in the radial, axial, and longitudinal directions, respectively.
For H<2R, the dominant mode of the circular cylindrical
cavity is transverse magnetic (TM010 )mode f3=fTM
010 ,
the cylindrical wall is added based on two considerations. One
is that it can improve the antenna performance on bandwidth
and forward radiation. The second is that it can be easily
expanded with lower interelement coupling.
In Case 3, no more resonance is added and f1is mov-
ing to a lower frequency at 2.05 GHz while f2is moving
to a higher frequency at 6.75 GHz. Two impedance band-
widths are increased slightly. Combining Case 2 and Case 3,
Case 4 demonstrates a fourth resonant frequency f4which
is created at 2.7 GHz since the transverse electric (TE111)
mode of the cavity is excited by the joint contribution of the
cylindrical electric wall and circular metallic plate. f4can be
calculated by f4=fTE
111 .Also, f1is at 1.9 GHz, f2is at
6.65 GHz, and f3is at 5.25 GHz. By employing the cavity,
two resonant frequencies are generated. Meantime, the cavity-
backed antenna raises the gain since the cavity’s metallic plate
can reflect the electromagnetic waves, acting as a reflective
surface like a mirror [20].
C. Evolution of Wideband Antenna
Fig. 5 illustrates the evolution of the proposed wideband
antenna where the switch is in the ON-state. Type I includes
a pair of mutually orthogonal dipole radiators (Antenna 1 and
Antenna 2) with a cavity. Type II is designed by applying
parasitic elements with shorted slots to Type I. Four bowtie-
shaped parasitic elements are arranged in the middle between
two radiators but locating on the other surface of the substrate.
The length of the parasitic element is about half the free-
space wavelength at the center frequency of the second band.
Type III is implemented by etching C-shaped slots to Type II.
Type IV is achieved by connecting vertical stubs in parallel
at feed ends of these radiators in Type III. The simulated
reflection coefficients and isolations of Types I, II, III, and
IV are shown in Fig. 6. |S22|and |S12|are similar to |S11|
and |S21|, respectively, due to rotational symmetry in space
[21]. By adding parasitic elements, the reflection coefficient
in the first band could perform better, which is observed
from the comparison between Type I and Type II in Fig. 6.
The first wideband feature is improved. Type I shows four
resonant frequencies, f1,f2,f3,and f4,where f1and f2are
kept close, and f3is adjacent to f4. When parasitic elements
with shorted slots are placed in the middle between both
radiating elements, the isolation |S21|between Antenna 1 and
Antenna 2 in Type II becomes weaker than that in Type I since
the couplings between radiating elements are increased. The
C-shaped slots in Type III can improve the second bandwidth
and enhance the isolation. In Type IV, the vertical stubs further
enhance the impedance matching of the second bandwidth.
The width of the vertical stub is equal to the width of
the feed end of the bowtie-shaped radiator. The height is
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7326 IEE E TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 69, NO. 11, NOVEMBER 2021
Fig. 5. Evolution of the proposed wideband antenna. (a) Type I.
(b) Type II. (c) Type III. (d) Type IV.
Fig. 6. Reflection coefficients and isolations of different antennas.
adjusted for the impedance matching. As can be seen from
Fig. 6, |S11|in the second band is improved, while other
parameters remain unchanged. The first and second bands of
|S11|<−10 dB are, respectively, from 1.58 to 2.78 GHz
and from 4.73 to 6.19 GHz, and the corresponding |S21|
are, respectively, larger than 41.7 and 36.2 dB. As can be
seen from Fig. 6, the vertical stub performs better in the
second band. Although such an assembly would make the
fabrication more complicated, this kind of operation provides
an essential reference for impedance matching considering:
1) a stub is directly connected in parallel while the total
size is not increased in such a cavity and 2) it is easy to
tune the height to realize impedance matching while other
physical parameters remain unchanged. From Fig. 6, there is
no difference between the basic structure and the final design
in terms of mutual coupling since the decoupling structure
of the interelement orthogonal arrangement is used in the
proposed wideband antenna. Afterward, the antenna design
is fine-tuned and finalized based on numerical simulations.
There was a slight difference between the initial and the final
values because of the inaccuracy of the design equations and
the existence of complex electromagnetic fields in the cavity.
D. Effects of C-Shaped Slots and Switch Slots
Both the C-shaped slots and switch slots can alter the
surface current distributions on the radiating elements and
parasitic elements, which further influence spatial electromag-
netic field distribution inside the cavity. The C-shaped slot is
initially positioned in the center of each radiator, and its length
is about half of the sum of the two ends of the trapezoid patch.
Fig. 7(a) and (b) illustrates the changes of lengths and offsets
of the C-shaped slots, where the arrow represents the direction
of the change under a certain variable. In Fig. 7(a), kand d
are kept unchanged while the slot length (S)is varied from
11.2, 13.2 to 15.2 mm, the simulated reflection coefficients
are shown in Fig. 7(c). It is found that as Sincreases, f1
and f4remain almost unchanged, while f2and f3move
simultaneously toward lower frequencies with the same second
bandwidth. The increase of the slot length increases the current
path along the radiator’s edge, thereby reducing high-order
Fig. 7. Effects of C-shaped slots on radiating elements. (a) Different slot
lengths. (b) Different slot offsets. (c) Reflection coefficients over slot lengths.
(d) Reflection coefficients over slot offset. (e) Frequency ratios of the second
band to the first band over dand S.
mode resonant frequency. Meantime, the frequency of TM
mode of the resonant cavity decreases since the increase of
the slot size reduces the electric/fields traveling back and forth
through the cavity. In Fig. 7(b), the C-shaped slot pattern is
kept unchanged, and the offset (d)is varied from 9.2, 7.2,
5.2, to 3.2 mm. From the simulated reflection coefficients
showninFig.7(d),asddecreases, f1,f2,and f4are almost
unchanged while f3moves toward a lower frequency, making
the second bandwidth wider. In the resonant cavity, the electric
fields gradually weaken from the center to the surrounding.
When the C-shaped slot is moving far away from the center,
the induced electric fields become smaller, and thereby f3is
reduced. With dand Svarying, the center frequency ratio of
the second band to the first band is illustrated in Fig. 7(e). As d
increases, the frequency ratio becomes smaller. As Sincreases,
the frequency ratio also becomes smaller. By changing dand
S, the frequency ratio can be tuned accurately from 2.3 to 2.74.
As illustrated in Fig. 2(c), the open slots on the parasitic
elements are investigated, which means that the switch is in
the OFF-state. The slot width ( p)is varied from 0.8, 1.0,
1.2 to 1.4 mm, and the corresponding reflection coefficients
are shown in Fig. 8(a). The results indicate that the second
bandwidth disappears. As pincreases, f1is moving toward
a lower frequency since the mutual capacitance between the
two patches of the parasitic element decreases [22]. Also, f4
remains unchanged, and the first bandwidth would be widened.
When the slot placement changes with the variable lfrom
22 to 1 mm, the reflection coefficients are simulated. From the
results shown in Fig. 8(b), four resonant frequencies are almost
unchanged. As ldecreases, the reflection coefficients close
to f2and f3are first getting higher and then getting lower.
When the slot is engraved close to the center of each parasitic
element, such as l=12 mm, the corresponding reflection
coefficient becomes worse. The open slots on the parasitic
elements can improve the first bandwidth and enable the close
of the second band. By controlling the switch, the switch slot
on the parasitic element can be operated as the shorted or open
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WANG et al.: COMPACT DUAL-POLARIZED WIDEBAND ANTENNA WITH DUAL-/SINGLE-BAND SHIFTING 7327
Fig. 8. Effects of open slots on parasitic elements. (a) Different slot widths.
(b) Different slot placements.
Fig. 9. Effects of the switch. (a) Switch offset from the center of the slot.
(b) Switch quantity on the slot. Reflection coefficients for (c) switch offset
from the center of the slot and (d) switch quantity on the slot.
slot, and therefore the proposed wideband antenna performs
the dual-band and single-band behaviors. The switch function
can be assisted by electricity, sound, light, and so on.
E. Effect of Switch Placement
When the switch is in the OFF-state, the parasitic element
is split into two parts, making the impedance matching of
the second band worsen. The influence is investigated on the
switch offset from the center of the slot and the switch quantity
on the slot, which are demonstrated in Fig. 9(a) and (b). In the
simulations, the switch is imitated with an internal resistance
of 0.02 m. When the switch offset wvaries from 0, 1,
and 3 mm, the reflection coefficients are shown in Fig. 9(c).
It is found that the resonances are almost unchanged, and the
magnitude in the second band has a little reduction. When
the switch quantity is gradually increased from 1, 2, and 3,
the reflection coefficients are shown in Fig. 9(d). It is found
that the resonances are almost unchanged, and the magnitudes
in these two bands are with a little reduction.
III. DIVERSITY PERFORMANCE ANALYSIS
In this section, several vital parameters for the proposed
wideband antenna are analyzed in terms of diversity gain
(DG), multiplexing efficiency, total active reflection coefficient
(TARC), channel capacity loss (CCL), and mean effective gain
(MEG). These diversity performance analyses could provide
Fig. 10. Calculated (a) ECC and DG, and (b) multiplexing efficiency.
essential references for massive MIMO antenna systems for
microbase station applications.
A. Diversity Gain
The envelope correlation coefficient (ECC) describes the
degree of independence or the correlation between antenna
elements at different frequencies. It can be calculated either
from the 3-D radiation patterns or from the S-parameters.
Given the antenna is lossless, the ECC between Antenna 1 and
Antenna 2 is calculated by
ECC =|S∗
11 S12 +S∗
21 S22|2
(1−|S11|2−|S21 |2)(1−|S22|2−|S12 |2).(7)
The calculated ECCs of the proposed wideband antenna are
displayed in Fig. 10(a). When the switch is in the ON-state,
the maximum values in the first and second bands are 1.74 ×
10−4and 1.04 ×10−3, respectively. When the switch is in
the OFF-state, the maximum value in the band of interest is
5.43 ×10−5.
The DG describes the performance of MIMO systems
against channel fading, and it is calculated by
DG=101−(ECC)2.(8)
The calculated results shown in Fig. 10(a) are high, nearly
10 dB in the operating bandwidth. A larger DG value indicates
better MIMO antenna performance. It can be observed that
the/proposed wideband antenna has the smaller ECC and the
larger DG simultaneously.
B. Multiplexing Efficiency
The multiplexing efficiency (ηmux )is calculated by
ηmux =(1−|ρc|2)η1η2(9)
where η1and η2are the total efficiencies of Antenna 1 and
Antenna 2, respectively, and ρcis the magnitude of the
complex correlation between Antenna 1 and Antenna 2, ρc=
√ECC. The calculated results are shown in Fig. 10(b). When
the switch is in the ON-state, ηmux is from −0.0468 to
−0.0323 dB in the first band and from −0.0724 to −0.041 dB
in the second band, respectively. When the switch is in the
OFF-state, ηmux is from −0.0539 to −0291 dB in the band of
interest. These values are larger than −3dB.
C. Total Active Reflection Coefficient
The TARC is defined as the ratio of the square root of total
reflected power divided by the square root of total incident
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7328 IEE E TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 69, NO. 11, NOVEMBER 2021
Fig. 11. Calculated (a) TARC and (b) CCL between antenna elements.
power. It can be calculated from S-parameters by
TARC =|(S11 +S12ejθ)|2+|(S21 +S22ejθ)|2
2(10)
where θrepresents the phase angle that swept from 0◦to 180◦.
The calculated TARC is shown in Fig. 11(a). It has stable
characteristics since the high isolation is observed between
Port 1 and Port 2 in the operating band.
D. Channel Capacity Loss
The CCL estimates the maximum limit up to which the
message transmission can occur without any loss in the com-
munication channel. At the high signal-to-noise ratio (SNR),
the CCL can be derived using S-parameters as
CCL =−log|ϕR|
2(11)
ϕR=1−|S11|2−|S12 |2−S∗
11 S12 +S∗
21 S22
−S∗
22 S21 +S∗
12 S111−|S22 |2−|S21|2.(12)
The calculated result is shown in Fig. 11(b). When the switch
is in the ON-state, the CCL values are 0.0145–0.2395 bits/s/Hz
in the first band and 0.0185–0.2257 bits/s/Hz in the second
band, respectively. When the switch is in the OFF-state, the
CCL values are 0.0124–0.2652 bits/s/Hz in the band of inter-
est. These values are much less than 0.4 bits/s/Hz, which is
the maximum acceptable value.
E. Mean Effective Gain
Considering environmental effects, the MEG is defined as
the ratio of power received by the diversity antenna to the
isotropic antenna’s power. For the proposed wideband antenna,
it can be calculated at Port 1 and Port 2, respectively, by
MEG1=1−|S11 |2−|S12|2
2(13)
MEG2=1−|S12 |2−|S22|2
2.(14)
The calculated results are shown in Fig. 12. When the switch
is in the ON-state, MEG1 is from −0.3783 to −0.3023 dB in
the first band and from −0.4116 to −0.3054 dB in the second
band, respectively; MEG2 is from −0.3837 to −0.3024 dB
in the first band and from −0.3966 to −0.3028 dB in the
second band, respectively. When the switch is in the OFF-
state, MEG1 and MEG2 are from −0.3898 to −3038 dB and
from −0.3888 to −0.3044 dB in bands of interest, respectively.
For the same power level, the ratios between MEG1 and
Fig. 12. Calculated MEG between antenna elements.
MEG2 are less than 0.005 dB, which meets the requirement
of less than 3 dB for a good MIMO antenna system. The
proposed wideband antenna could offer MIMO capability.
After the replication, the expanded antenna array of four and
more elements can be obtained with low mutual coupling,
which will further enhance MIMO capability.
IV. FABRICATION AND MEASUREMENT
A. Prototype Fabrication and Measurement Setup
To verify the feasibility of the design concept, an expanded
four-element MIMO antenna prototype of the proposed wide-
band antenna is manufactured by the manual assembly, and
an external associated circuit realizes its switch function.
The configuration of the fabricated prototype is illustrated
in Fig. 13. The separation distance between two adjacent
proposed wideband antennas can be zero, that is, the center
distance (D)is 106 mm, about three-quarters at the center
frequency of the first band. Fig. 13(a) shows the support
of the cavity, which is printed by using the 3-D printing
technology. The copper foil with a thickness of 0.0356 mm is
spread inside it, as shown in Fig. 13(b). Then, two semirigid
coaxial cables with a characteristic impedance of 50 are
stick out from the bottom, with a pair of connector structures
connecting with radiating elements, as shown in Fig. 13(c).
In Fig. 13(d), the parasitic elements with soldered wires
and the assembled/vertical stubs are vertically fixed through
vias, further soldered together. The switch function is realized
through four relays, a microcomputer unit (MCU) and a
manual button, where the relays are connected to parasitic
elements with wires and controlled by the MCU. By pressing
on the button, the four relays can work simultaneously, and
then the proposed wideband antenna is imitated to operate in
the ON-state or OFF-state, as shown in Fig. 13(e). Fig. 13(f)
shows an expanded four-element MIMO antenna prototype.
The measurement environment is shown in Fig. 13(g). The
measurements are carried out for the fabricated prototype in
an anechoic chamber equipped with an Agilent vector network
analyzer N5230A, two identical broadband horn antennas LB-
8180-SF, and the far-field measurement system.
B. Measured and Simulated Results
As shown in Fig. 14, the measured and simulated reflec-
tion coefficients and interelement isolations are acquired and
compared. When the switch is in the ON-state, two frequency
bands are created since two parts of each parasitic element
are connected to form larger induced current paths. The first
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WANG et al.: COMPACT DUAL-POLARIZED WIDEBAND ANTENNA WITH DUAL-/SINGLE-BAND SHIFTING 7329
Fig. 13. Expanded four-element MIMO antenna prototype of the proposed
wideband antenna for the proof of concept and its measurement. (a) 3-D
printing support of the cavity. (b) Spread copper foil inside the support.
(c) Coaxial cables and connector structures. (d) Parasitic elements and vertical
stubs. (e) Fabricated wideband MIMO antenna and its associated circuit.
(f) Expanded four-element MIMO antenna prototype. (g) Measurement setup
and its equipment.
bandwidth covers the range of 1.58–2.77 GHz, and the second
bandwidth covers the range of 4.71–6.18 GHz. When the
switchisintheOFF-state, the parasitic elements are divided
into two separate parts so that the magnitude of high-frequency
reflection coefficients becomes larger than −6 dB, which is not
acceptable in wireless communications [23]. Hence, the dual-
band is shifted into the single-band. It covers the range of
1.56–2.76 GHz, and the overlapped bandwidth is from 1.58 to
2.76 GHz regardless of whether the switch is ON or OFF-state.
From Fig. 14, it should be noted that all measured isolations
in the bands of interest are more than 35.3 dB. As shown
in Fig. 15(a), when the switch is in the ON-state or OFF-
state, all measured and simulated ECCs are estimated less
than 0.002 in the wide frequency bands. Fig. 15(b) shows
the measured and simulated radiation efficiency and realized
gain considering losses of coaxial cables and SMA connectors.
Single-band and dual-band are shifted by controlling the
switch. When the switch is in the ON-state, the measured
radiation efficiency varies in the range of 87.5%–91% in the
first band and 87%–91% in the second band; the measured
realized gain varies in the ranges of 9.4–11.2 dBi in the first
band and 9.1–10.9 dBi in the second band. When the switch
is in the OFF-state, the measured radiation efficiency varies in
the range of 88%–92%; the measured realized gain varies in
the range of 8.9–10.1 dBi. The radiation efficiency is falling
in the second band as the currents on radiators are widely
distributed, which could be reflected by the side lobes and
high cross-polarizations in the radiation patterns.
The proposed wideband antenna can achieve polarization
diversity, including two mutually perpendicular linear polar-
izations. The linear polarization of Antenna 1 is perpendic-
ular to the linear polarization of Antenna 2 in both bands.
Fig. 14. Measured and simulated reflection coefficients and isolations with
(a) ON-state dual-band and (b) OFF-state single-band.
Fig. 15. Measured and simulated (a) ECCs and (b) radiation efficiency and
realized gain.
By manually pressing the button to start the ON-state or
OFF-state for shifting between the dual-band and single-
band, the radiation patterns in the x
z
plane and y
z
plane
are measured in the fully calibrated anechoic chamber with
an automated far-field measurement system. The measured
frequencies are at 2.4 and 5.8 GHz, and the results are taken
by driving one antenna element and terminating others to loads
of 50 , as shown in Figs. 16 and 17. It is observed that they
are unidirectional radiations. The measured cross-polarization
levels are lower than −19.3 dB in both the x
z
plane and y
z
plane. Moreover, the measured front-to-back ratio, which is
the ratio of power radiated in the broadside radiation and
the power radiated in the opposite direction, remains over
20 dB. When the switch is in the ON-state or OFF-state,
the co-polarization patterns at 2.4 GHz are nearly the same,
and correspondingly, the cross polarization becomes small,
less than −20 dB. However, the cross-polarizations in the
OFF-state become much smaller than these in the ON-state.
When the proposed wideband antenna operates at 5.8 GHz,
it is apparently to find that the radiation pattern in the x
z
plane
significantly differs from that in the y
z
plane shown in Fig. 16,
which is mainly caused by asymmetrical feed structure, via
perturbation, and high-order resonance mode [2], [14], [24].
Comparing the measured results with the simulated results,
several slight differences at frequency points or certain angles
are associated with manufactured proof-of-concept prototype’s
imperfections. Although the presented switch seems to yield
additional cost, it raises an approach to meet this kind of
requirement. More cost-effective functioned chips as switches
would be explored and controlled by sound, light, and so on.
C. Comparison With Other Published Works
The proposed wideband antenna is compared with the
published works as summarized in Table I. They all work in
the microwave band using two input ports. Although these
designs in [6], [25], and [26] achieve the dual bands, they are
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7330 IEE E TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 69, NO. 11, NOVEMBER 2021
TAB L E I
COMPARISON OF THE PROPOSED WIDEBAND ANTENNA AND OTHER PUBLISHED WORKS
Fig. 16. Measured and simulated ON-state radiation patterns at (a) 2.4 and
(b) 5.8 GHz.
single-polarized radiation and the SNR from vertical directions
following this polarization is very low. Also, the complex
decoupling structures are designed to improve the isolation
in [6] and [25]. Orthogonal placement improves isolation
significantly from the physical structure itself without adding
additional design and their isolation levels can be larger than
23 dB [16], [21], [24], [27]–[31]. Also, polarization diversity
can be achieved. Most of the designs only work in a single
wide bandwidth (>20%). Although the design in [31] has
the largest FBW, its height is relatively thicker. The designs
in [24] and [32] realize the dual-polarized and dual-band
performance; however, the bandwidths are relatively narrower
(<20%). Besides, the height in [24] is relatively large.
In comparison with these designs, the proposed wideband
antenna not only owns the ability to obtain better radiation
performances across the two bands but achieves a shifting
between dual-band and single-band. Although the design
in [26] could shift in two bands, the gain in the first band
is low, and at the same time, the polarization is single
linear-polarization. Furthermore, the realized gain of the
proposed wideband antenna maintains an excellent property
across the two desired frequency bands, which demonstrates
the feasibility of the design methodology. This design
methodology could be repeated to design similar antennas
working at different bands.
The proposed wideband antenna is compact and easy to
fabricate. Moreover, the separation between the two frequency
bands can be flexibly adjusted by taking advantage of the
Fig. 17. Measured and simulated OFF-state radiation patterns at 2.4 GHz.
C-shaped slot of the radiator in a suitable range. The working
principle of the proposed wideband antenna needs the cooper-
ative operation of both the cavity and radiators. Although the
whole antenna seems to be bulky, it provides better unidirec-
tional radiation and reduces the backward lobes. The cavity
not only greatly reduces the impact of the installation place
on the antenna performance but also reduces the influence of
radiators on the adjacent antennas, maintaining stable antenna
performance.
D. MIMO System Experiment
The proposed wideband antenna can operate as either the
transmitter antenna or receiver antenna in the MIMO system.
The antenna offers two ports: one for vertical radiation and the
other for horizontal radiation in broadside mode. Two ports
could receive/transmit mutually orthogonal linear-polarization
signals, and the interference could be greatly reduced. As the
receiver antenna, only one port would receive most of the
transmitted signal of the same polarization and obtain a good
SNR, while the other would obtain a poor SNR.
A2×2 MIMO system is set up as shown in Fig. 18(a). One
of the proposed wideband antennas is used as the transmitter
antenna, the other as receiver antenna. The MIMO system
uses NI LabVIEW software and NI USRP hardware with
Alamouti space-time block coding and maximal ratio com-
bining. The carrier frequencies are set to be 2.5 and 3.5 GHz,
respectively, and the modulation scheme is 4-QAM. The same
data stream is transmitted through different antennas and then
combined at the receiver side. When the carrier frequency is at
2.5 GHz, the tested bit error rate (BER) performance is shown
in Fig. 18(b). As SNR increases, the BER is exponentially
reducing. When the BER is 1.0×10−4,Eb/N0is 15.51 dB.
By comparison, only Antenna 1 or Antenna 2 is utilized to
form single input single output (SISO) system at 2.5 GHz.
The BER curve has the same trend, but it is slightly higher
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WANG et al.: COMPACT DUAL-POLARIZED WIDEBAND ANTENNA WITH DUAL-/SINGLE-BAND SHIFTING 7331
Fig. 18. (a) Experimental 2 ×2 MIMO system. (b) Tested BER over Eb/N0.
than that of an MIMO system. To achieve the same BER
level, Eb/N0needs to be 18.55 dB. Similarly, at 3.5 GHz,
the same phenomena are observed. The 2 ×2MIMOsystem
has a lower BER than that of the SISO system with only
Antenna 1/Antenna 2, enhancing the reliability of transmission
and link. The proposed wideband antenna is fairly good for
typical microbase station applications in IoV, covering the
frequency bands of WLAN (2.4–2.48 GHz, 5.15–5.925 GHz),
Wi-Fi (2.4–2.4835 GHz, 5.150–5.350 GHz, 5.725–5.825 GHz,
5.470–5.725 GHz), LTE (bands 1–4.7, 9, 10, 23, 25, 30,
33–41, 46, 47, 53, 65, 66, 69, 70, 252, 255), Universal Mobile
Telecommunications Service (UMTS) (1.92–2.17 GHz), and
personal communication services (PCS) (1.85–1.99 GHz).
V. C ONCLUSION
In this article, a novel compact wideband MIMO antenna
with dual-polarized, high-isolated, and high-gain features has
been presented for microbase stations in IoV. It can shift
between the dual-band and single-band. Theoretical analysis
and experimental verification are conducted. Four resonant
frequencies are generated by the joint contribution of the
radiators and cavity, which further forms dual-wideband band-
width. The C-shaped slots on radiators and switch slots on
parasitic elements are analyzed for estimating the impact on
resonant frequencies and bandwidths. Orthogonal arrangement
significantly reduces mutual decoupling, and the addition of
vertical stubs in parallel improves impedance matching.
As a proof of concept, an expanded four-element array
of the proposed wideband antenna is prototyped. An exter-
nal associated circuit initially attains the switch function.
Both measured and simulated results consistently validate the
diversity-enhanced performance. When the switch is in the
ON-state, it operates at 2.175 and 5.445 GHz with bandwidths
of 1.19 and 1.47 GHz, peak radiation efficiencies of 91%
and 91%, and peak realized gains of 11.2 and 10.9 dBi,
respectively. When the switch is in the OFF-state, it operates
at 2.16 GHz with a bandwidth of 1.2 GHz, peak radiation
efficiency of 92%, and the peak realized gain of 10.1 dBi.
Moreover, element isolation of more than 35.3 dB and ECCs
of lower than 0.002 are achieved within the bands. From the
experiments of the established 2 ×2 MIMO system, the BER
is lower than that of the SISO system, and the proposed
wideband antenna is an ideal candidate for massive MIMO
microbase station antennas.
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Wensong Wang (Member, IEEE) received the Ph.D.
degree from the Nanjing University of Aeronautics
and Astronautics, Nanjing, China, in 2016.
From 2013 to 2015, he was a Visiting Scholar
with the University of South Carolina, Columbia,
SC, USA. In 2017, he joined Nanyang Technological
University, Singapore, as a Research Fellow. His
research interests include RF/microwave systems,
inter-/intrachip wireless interconnect, power wireless
transfer, signal integrity, and nondestructive real-
time health monitoring.
Shuhui Yang (Member, IEEE) received the B.Sc.
degree from Zhejiang University, Hangzhou, China,
in 1994, the M.Sc. degree from the Communication
University of China, Beijing, China, in 2000, and the
Ph.D. degree from The Institute of Microelectron-
ics of the Chinese Academy of Sciences, Beijing,
in 2003.
From 2003 to 2015, he was a Professor with
Beijing Information Science & Technology Univer-
sity, Beijing. In 2008, he was a Visiting Scholar
and an Adjunct Associate Professor with the Uni-
versity of South Carolina (USC), Columbia, SC, USA. In 2011, he joined
Victoria University, Melbourne, VIC, Australia, as a Visiting Scholar. From
2013 to 2014, he was a Visiting Professor with USC. In 2015, he joined
the Communication University of China, where he is currently a Professor
and the Director of the Department of Communication Engineering. His
current research interests include the electromagnetic metamaterials, passive
microwave components, electromagnetic compatibility (EMC), signal integrity
(SI), and wireless inter-/intrachip communication system.
Dr. Yang is a Senior Member of the Chinese Institute of Electronics (CIE).
Zhongyuan Fang (Member, IEEE) received the
B.Sc. degree in microelectronics from Fudan Univer-
sity, Shanghai, China, in 2016, and the Ph.D. degree
from Nanyang Technological University, Singapore,
in 2021.
His research interests include analog/mixed-signal
integrated circuit design, energy-efficient algorithms
for physiological signal processing, low-power sen-
sor interface IC design, as well as RF/wireless com-
munication circuits and systems design and testing
for portable biomedical applications.
Quqin Sun received the B.Sc. and Ph.D. degrees
from the Huazhong University of Science and
Technology, Wuhan, China, in 2011 and 2016,
respectively.
He worked as a Research Associate with the
Institute of Fluid Physics, China Academy of
Engineering Physics, Mianyang, China, till 2018.
He is currently a Research Fellow with Nanyang
Technological University, Singapore. His current
research interests include electromagnetic coil
design, machine learning, and nondestructive testing.
Yinchao Chen (Senior Member, IEEE) received
the Ph.D. degree in electrical engineering from the
University of South Carolina (USC), Columbia, SC,
USA, in 1992.
Since then, he has worked with the University
of Illinois at Urbana–Champaign (UIUC), Urbana,
IL, USA, The Hong Kong Polytechnic University
(HKPU), Hong Kong, and USC, respectively. He is
one of the coauthors or editors for three academic
books and ten book chapters in electrical engi-
neering, and has authored more than 200 academic
articles in international journals and conference proceedings. His current
research interests include signal integrity for high-speed circuits, RF and
integrated microwave circuits, and integrated microstrip antennas.
Yuanjin Zheng (Senior Member, IEEE) received
the B.Eng. and M.Eng. degrees from Xi’an Jiaotong
University, Xi’an, China, in 1993 and 1996, respec-
tively, and the Ph.D. degree from Nanyang Techno-
logical University, Singapore, in 2001.
He was with the National Key Laboratory of Opti-
cal Communication Technology, University of Elec-
tronic Science and Technology of China, Chengdu,
China, from 1996 to 1998. In 2001, he joined the
Institute of Microelectronics (IME), Agency for Sci-
ence, Technology and Research, Singapore, where
he was a Principal Investigator and the Group Leader. In 2009, he joined
Nanyang Technological University, and is currently the Center Director of
the VIRTUS IC Design Center of Excellence and the Program Director
of VALENS Bio Instrumentation, Devices & Signal Processing. With IME,
he led and developed various projects like CMOS RF transceivers, baseband
system-on-chip (SoC) for wireless systems, ultra-wideband, and low-power
biomedical ICs. He has authored or coauthored over 400 international journal
articles and conference papers, and several book chapters, and holds 26 patents
filed/granted. His current research interests include gigahertz radio frequency
integrated circuit and SoC design, biosensors and imaging, and surface
acoustic wave/bulk acoustic wave/micro-electromechanical systems sensors.
Dr. Zheng has been organizing over 15 conferences as a TPC and Session
Chair, and has delivered over 25 invited talks at international conferences.
He was an Associate Editor of IE EE TRANSACTIONS ON BIOMEDICAL
CIRCUITS AND SYSTEMS, and currently serves as an Associate Editor
for IEEE JOURNAL OF ELECTROMAGNETICS,RF AND MICROWAVE IN
MEDICINE AND BIOLOGY (J-ERM).
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