A Planar End-Fire Circularly Polarized Complementary Antenna With Beam in Parallel With Its Plane

Abstract

Operation principle and design approach of a novel planar end-fire circularly polarized (CP) complementary antenna is proposed. A vertically polarized printed magnetic dipole and a horizontally polarized printed dipole are combined on the same substrate, and a planar CP antenna with end-fire beam in parallel with its plane is thus designed. Prototype antennas centered at 5.80 GHz are then fabricated and measured to validate the operation principle and the design approach. The experimental prototype reported that the impedance bandwidth (20 log |S11| < - 10 dB) is about 1.90%, from 5.75 to 5.86 GHz and the 3-dB axial ratio (AR) bandwidth is about 14.48%, from 5.19 to 6.00 GHz. Therefore, the proposed design is applicable as a low-profile handheld reader antenna in radio-frequency identification (RFID) systems.

Full text

1146 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 3, MARCH 2016
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A Planar End-Fire Circularly Polarized Complementary
Antenna With Beam in Parallel With Its Plane
Wen-Hai Zhang, Wen-Jun Lu, and Kam-Weng Tam
Abstract—Operation principle and design approach of a novel pla-
nar end-fire circularly polarized (CP) complementary antenna is pro-
posed. A vertically polarized printed magnetic dipole and a horizontally
polarized printed dipole are combined on the same substrate, and a pla-
nar CP antenna with end-fire beam in parallel with its plane is thus
designed. Prototype antennas centered at 5.80 GHz are then fabricated
and measured to validate the operation principle and the design approach.
The experimental prototype reported that the impedance bandwidth
(20 log |S11|<−10 dB) is about 1.90%, from 5.75 to 5.86 GHz and the
3-dB axial ratio (AR) bandwidth is about 14.48%, from 5.19 to 6.00 GHz.
Therefore, the proposed design is applicable as a low-profile handheld
reader antenna in radio-frequency identification (RFID) systems.
Index Terms—Circularly polarized (CP), complementary antenna,
end fire, handheld radio-frequency identification (RFID) reader, planar
antenna.
I. INTRODUCTION
The circularly polarized (CP) antennas have been intensively stud-
ied since 1940s [1], [2]. The design approaches to generate CP radia-
tion can be basically categorized into five distinctive types, according
to their operation principles. The first type is to use the superposi-
tion of complementary dipoles [1]–[5], such as the combination of
dipole/monopole and loop [1], slotted cylindrical dipoles [3], and
printed strips and slots [4]. Generally, both broadside [1]–[4] and end-
fire [5] CP radiation beams can be realized by using these techniques.
The second one is to use the superposition of identically orthogonal
dipoles. The use of two orthogonally crossed dipole elements fed with
equal magnitude and 90◦phase difference is a straightforward way
to generate broadside, CP radiations [6]–[9]. The third way to gen-
erate CP radiation is to introduce the turnstile structures, including
the helices [10], [11] and the spirals [12]–[15]. In this case, end-fire
CP radiation beams can be obtained by using relatively electrically
large turnstile structures [10]–[15]. With reference to planar spirals or
helices configuration, the end-fire beam is perpendicular to the plane
of the antenna [13]–[15]. The forth type is to excite two degenerate
modes within a single radiator, e.g., a patch [16], [17] or a dielectric
resonator [18], [19], by employing 90◦hybrid couplers/dividers [16] or
using perturbations [17]–[19]. As similar to the previously described
three distinctive types of CP antenna design methods, the resulted radi-
ation patterns of this method are also broadside. The final one utilizes
the traveling-wave or periodical structures, such as the waveguide-fed
horn antennas [20], [21], the substrate integrated waveguide antennas
Manuscript received June 24, 2015; revised November 22, 2015; accepted
January 03, 2016. Date of publication January 14, 2016; date of current ver-
sion March 01, 2016. This work was supported in part by FDCT project
015/2013/A1, One-Time Special Fund for Ph.D. Student Support of the
University of Macau and National Natural Science Foundation of China under
Grant 61471204.
W.-H. Zhang and K.-W. Tam are with the Department of Electrical and
Computer Engineering, Faculty of Science and Technology, University of
Macau, Macao, China (e-mail: yb47413@umac.mo; kentam@umac.mo).
W.-J. Lu is with the Jiangsu Key Laboratory of Wireless Communications,
Nanjing University of Posts and Telecommunications, Nanjing 21003, China
(e-mail: wjlu@njupt.edu.cn).
Color versions of one or more of the figures in this communication are
available online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2016.2518204
0018-926X © 2016 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 3, MARCH 2016 1147
[22]–[25], and the metamaterial-based leaky wave antennas [26], [27].
Most of them are electrically large and exhibit broadside CP beams.
By comparing the five types of CP antenna design approaches accord-
ing to their basic operation mechanisms and radiation behaviors, it is
found that to design a planar CP antenna with an end-fire beam in
parallel with its plane while keeping a planar antenna structure, is a
challenging task. Although an end-fire CP beam in parallel with the
major plane of the antenna can be realized [25], [27], nonplanar con-
figurations will be introduced due to the inherent feed structure [5] or
introducing pairs of additional orthogonal elements [25], [27].
On the other hand, due to broadside radiation characteristic, the
antennas are often assembled perpendicularly to the reader to achieve
front-directional radiation for handheld radio-frequency identification
(RFID) readers, which significantly increases the whole profile of
handheld reader [28], [29]. Recently, a new approach to design planar,
end-fire CP antenna having a beam in parallel with its plane is studied
[30]. Different from combining a pair of orthogonally magnetic dipole
together [30], a new planar complementary antenna with a simpler
structure is proposed in this communication. The proposed antenna
is combined with an aperture and a printed dipole. To the best of our
knowledge, this is the first time using the complementary dipoles to
achieve a planar end-fire CP antenna with its beam in parallel with the
plane. This communication is organized as follows. In Section II, an
equivalent sources’ model is presented to deduce the dipole-aperture
combined antenna configuration, and the design guideline is addressed
as well. In Section III, the proposed antenna is parametrically stud-
ied and optimal parameters are obtained. In Section IV, prototypes are
fabricated, measured, and compared to validate the design approach.
II. THEORY AND DESIGN GUIDELINE
Conceptual configuration consisting of a pair of paralleled magnetic
and electric dipole is shown in Fig. 1(a), which aligns with the y-axis.
The aperture element is equivalent to a virtual time-harmonic magnetic
dipole. The electric field at an arbitrary observation point Pin the far-
field can be obtained, according to [31]

Eaperture =
Eθ+
Eϕ=jωμ0I0l
4πηr ˆ
θcos ϕ+ˆϕcos θsin ϕe−jkr
(1)
where I0is the amplitude, lis the length of dipole, η=μ0/ε0is the
wave impedance, μ0and ε0are the permittivity and the permeability
of free space, respectively.
On the other hand, the far-field pattern of y-oriented electric dipole
is [31], [32]

Edipole =
Eθ+
Eϕ=jωμ0I0l
4πηr ˆ
θcos θsin ϕ+ˆϕcos ϕe−jkr.
(2)
When the aperture and printed electric dipole are excited with equal
amplitude with a distance of dabout 3λ/8(λis the guided-wave
wavelength and k=2π/λ is the wavenumber). Here, δ0indicates the
temporal phase that is caused by the current flowing from the aperture
to the printed dipole, and it is equal to kd =3π/4. The total far field
of the complementary configuration will be

Etotal =
Eaperture +e−j(kd sin θcos ϕ+δ0)
Edipole
=jωμ0I0l
4πηr ˆ
θ(cos ϕ+cosθsin ϕ(f(θ, ϕ)))
+ˆϕ(cos θsin ϕ+cosϕ(f(θ, ϕ)))e−jkr
(3)
Fig. 1. Planar end-fire complementary CP antenna. (a) Equivalent sources’
model. (b) Conceptual design. (c) Sectional view of the proposed antenna.
where
f(θ, ϕ)=cos
3π
4sin θcos ϕ+3π
4−jsin 3π
4sin θcos ϕ+3π
4.
(4)
When θ=90
◦,ϕ=0
◦, the E-field along the +x-axis is

E+x=jωμ0I0l
4πηr ˆ
θ+ˆϕje−jkr.(5)
From (5), it is seen perfect CP radiation characteristic can be
obtained at the +x-direction, i.e., the end-fire direction of the planar
aperture-dipole combined antenna.
To more clearly reveal the mechanism for CP generation of the
proposed antenna, radiation patterns of a single microstrip magnetic
dipole at the resonant frequency are simulated in Fig. 2. It is observed
that the antenna produces an “eight”-shaped radiation pattern in the
azimuth plane (xy-plane and H-plane), while an “O”-shaped radiation
pattern in the elevation plane (xz-plane and E-plane). The result proves
that the antenna resonates as a vertically polarized (i.e., z-polarized)
magnetic dipole operating at its dominant, one-half wavelength mode.

1148 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 3, MARCH 2016
Fig. 2. Simulated radiation patterns at 5.8 GHz. (a) xy-plane. (b) xz-plane.
Fig. 3. Geometry of the proposed antenna.
It exhibits radiation maxima at the end-fire +x-direction [33]. For a y-
polarized electric dipole operating at its dominant mode, it should have
a complementary radiation pattern and orthogonal polarization with
the magnetic dipole. Therefore, if 90◦phase difference is provided,
end-fire CP characteristic can be achieved while a planar structure is
maintained.
Based on the theoretical and numerical analysis, the basic structure
of the planar CP complementary antenna can be deduced. In Fig. 1(b)
and (c), three edges of magnetic microstrip dipole are shorted with one
left opened [33] to form an aperture. The aperture serves as a virtual
magnetic microstrip dipole that operates at one-half wavelength mode
with width of one-quarter wavelength [34]. A coaxial cable probe is
used to excite the magnetic dipole and is located near its E-current’s
antinode [30]. A pair of 3λ/8, broadside-coupled stripline [35] is
placed between the aperture and printed electric dipole to introduce
a proper phase difference. In this way, a planar end-fire CP beam in
parallel with the substrate’s plane will be resulted in.
III. ANTENNA GEOMETRY AND PARAMETRIC STUDY
The geometry of the proposed antenna is shown in Fig. 3. The pro-
posed antenna with a total size of 38 mm ×33.5 mm is designed on
the substrate with a relative permittivity of εr=2.65,tanδ=0.001,
and thickness h=2.0mm.
The direction of CP of the proposed antenna can be determined by
the probe current’s path/loop [30]. Suppose the probe is fed to the top
layer of the magnetic dipole, and the current flows from the probe, then
to the top layer of the broadside-coupled stripline, the top arm of the
printed dipole. To maintain the continuity of the current path/loop, it is
supposed that the displacement current flows to the +x-direction, and
TAB L E I
ANTENNA PARAMETERS
Fig. 4. Effect of the angle of magnetic dipole α.
Fig. 5. Effect of the distance of feed point d.
then turns back to the bottom arm printed dipole, the bottom layer of
the broadside-coupled stripline, and finally flows into the bottom layer
of the magnetic dipole. If such a virtual probe current’s path obeys
a right-handed helix direction with thumb point at +z-direction, a
right-handed CP (RHCP) antenna is obtained. Otherwise, a left-handed
CP (LHCP) one is resulted instead. In this communication, an RHCP
antenna is designed and investigated.
In order to increase the bandwidth, a printed fatter dipole is adopted
[36]. The initial parameters are determined according to the design
guideline and the empirical results in [30] and [33], and tabulated in
Table I. In order to simultaneously achieve the optimized impedance
bandwidth and axial ratio (AR) characteristics, the parametric studies
are carried out to determine the optimal parameters. The antenna is
modeled by using Zeland’s IE3D v12.0 with method-of-moment codes
and finite substrate is considered in simulations. When one parameter
is studied, the others are kept unchanged.
The effect of angle αon the antenna performance is studied first in
Fig. 4. It is observed that it has a significant effect on impedance char-
acteristic, the center frequency shifts up for a larger α.Fig.5shows
the effect of the distance of feed point. As we can see better impedance
characteristic can be achieved when dis equal to 2 mm. However, both
of them have slight effect on 3-dB AR bandwidth.
Fig. 6 shows the effect of length of broadside-coupled stripline on
the performance of the proposed antenna. It is observed that with the

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 3, MARCH 2016 1149
Fig. 6. Effect of the length of broadside-coupled stripline L1.
Fig. 7. Effect of the width of printed electric dipole L2.
Fig. 8. Effect of the length of printed electric dipole W2.
Fig. 9. Surface current distribution at 5.8 GHz.
Fig. 10. Top- and bottom-view photographs of the prototype. (a) Top view.
(b) Bottom view.
Fig. 11. Measured and simulated reflection coefficients of the proposed
antenna.
Fig. 12. Measured and simulated ARs at +x-direction (θ=90
◦and ϕ=0
◦)
of the proposed antenna.
Fig. 13. Measured and simulated gains of the proposed antenna.

1150 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 3, MARCH 2016
Fig. 14. Measured and simulated efficiencies of the proposed antenna.
increasing L1, it improves 3-dB bandwidth significantly and the mini-
mized AR shifts up gradually. The effect of the width and the length of
the printed dipole are also studied. In Fig. 7, with the increasing L2,the
minimized AR shifts down and better 3-dB AR bandwidth is achieved
when L2is equal to 6 mm. However, If L1and L2decrease contin-
uously, AR will become worse. Fig. 8 shows that when the length
increases, 3-dB AR bandwidth becomes much wider and minimized
AR shifts down gradually.
In order to show the mechanism for CP generation, simulated sur-
face current distributions at 5.8 GHz are investigated and shown in
Fig. 9. It is seen that both the magnetic and electric dipoles are operat-
ing at their dominant mode, as theoretically predicted in the previous
section. When a 90◦phase difference is properly introduced, end-fire
CP radiation characteristic can be achieved.
IV. SIMULATED AND EXPERIMENTAL RESULTS
Based on the parametric studies, prototypes of the proposed
antenna, shown in Fig. 10, are fabricated and measured to verify
the design approach. The reflection coefficient is measured by using
an Agilent N5230A vector network analyzer. The radiation patterns,
gain, efficiency, and AR are measured in a Satimo’s Starlab near-field
antenna measurement system.
Fig. 11 shows the measured and simulated reflection coefficients
of the proposed antenna. The measured −10-dB reflection coefficient
bandwidth is from 5.75 to 5.86 GHz, about 1.90% in fractional, while
the simulated one is from 5.70 to 5.91 GHz, about 3.62% in frac-
tional. The mismatch is caused by the fabrication and measurement
errors caused by the nonideal probe and the SMA connector. Fig. 12
shows the measured and simulated 3-dB AR bandwidth of the pro-
posed antenna. The measured AR bandwidth is from 5.18 to 6.00 GHz,
which is larger than 14.48% in fractional, while the simulated one is
from 5.18 to 5.98 GHz, which is about 14.34%. Both of them are wider
than the corresponding impedance bandwidth. The measured and sim-
ulated gains at +x-direction and efficiency of the proposed antenna are
illustrated in Figs. 13 and 14, respectively. It is observed that the pro-
posed antenna exhibits stable gain at +x-direction of about 2.3 dBic
within its impedance bandwidth. The measured gain is in good agree-
ment with the simulated one in the low-frequency regime and some
discrepancies can be observed in the high-frequency band, i.e., above
5.80 GHz. The average measured efficiency is about 78%, slightly
lower than the simulated one. The measured and simulated results,
including the reflection coefficient, AR, gain, and efficiency, are in
good agreement with each other.
The measured and simulated radiation patterns in xz-plane and xy-
plane at 5.65 GHz (the minimized AR) and 5.80 GHz (the center
Fig. 15. Measured and simulated radiation patterns at 5.65 GHz. (a) xz-plane.
(b) xy-plane.
Fig. 16. Measured and simulated radiation patterns at 5.80 GHz. (a) xz-plane.
(b) xy-plane.
TAB L E I I
BEAMWIDTHS OF THE ANTENNA
frequency of impedance band) are plotted in Figs. 15 and 16, respec-
tively. It can be observed that the simulated results are consistent
with the measured ones. The proposed antenna has a +x-direction,
end-fire CP radiation pattern, as predicted in the above. In both
principal planes, symmetrical radiation patterns and wide angle AR
characteristics are obtained. The measured half-power beamwidths
of the proposed antenna are 155◦and 75◦in xz-plane and xy-plane,
respectively, while the corresponding 3-dB AR beamwidth is 110◦and
80◦at 5.80 GHz, which is desirable for wide-coverage RFID appli-
cations [37]. The measured and simulated half-power and 3-dB AR
beamwidths of both frequencies are summarized in Table II.
A comprehensive comparison with other typical CP antennas is
also presented. The comparisons in terms of operation principles,
beam directions, geometry, and 3-dB AR bandwidth are tabulated
in Table III. It is seen that most planar antennas can only achieve
CP beam, which is perpendicular to the plane except [30], and the
new complementary antenna has an end-fire CP beam in parallel
with its plane. Furthermore, the CP operation bandwidth is increased
to 14.48% compared to its planar counterpart using orthogonally
combined magnetic dipoles [30]. This is possibly caused by the
mutual cancellation of reactance characteristics of the complementary
dipoles [38].

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 3, MARCH 2016 1151
TABLE III
FIGURE OF MERIT COMPARISONS
V. CONCLUSION
In this work, a novel planar dipole-aperture combined antenna has
been proposed, fabricated, and tested. An end-fire CP beam in parallel
with its substrate plane is obtained. Our study shows that the proposed
antenna can achieve an impedance bandwidth of 1.90% and a 3-dB AR
bandwidth of 14.48%. The measured results exhibit a good agreement
with the predicted ones. The principle of end-fire CP beam realiza-
tion has been demonstrated and experimentally validated. Therefore,
the proposed antenna is advantageous for low-profile handheld RFID
reader applications.
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Antenna Measurements in Reverberation Chamber
Using USRP
Ahmed Hussain, Andrés Alayón Glazunov, Bjarni Þór Einarsson,
and Per-Simon Kildal
Abstract—This paper shows that a universal software radio periph-
eral (USRP) can be used to measure the complete passive over-the-air
(OTA) performance of multiport antennas for multiple-input multiple-
output (MIMO) systems. The passive MIMO OTA measurements are
performed in a reverberation chamber using the USRP as a measurement
instrument. The total embedded radiation efficiencies, the correlation,
and the resulting diversity gain are extracted from transmission measure-
ments only. These are in good agreement with S-parameter measurements
(both transmission and reflection) obtained with a vector network analyzer
(VNA). The proposed measurement approach involving USRPs provide
an economical alternative to these type of measurements as compared to
high-end equipment.
Index Terms—Correlation coefficient, diversity gain, efficiency,
multiple-input multiple-output (MIMO), over-the-air (OTA), rever-
beration chamber (RC), universal software radio peripheral
(USRP).
I. INTRODUCTION
The latest wireless communication technologies require more
expensive testing and measurement equipment to accurately charac-
terize the performance of the increasingly complex wireless devices.
A less expensive alternative can be found in the universal software
radio peripheral (USRP), which is an active, programmable device [1].
USRPs have been proved to be useful for complete active multiple-
input multiple-output (MIMO) over-the-air (OTA) characterization in
terms of throughput, total isotropic sensitivity, and total radiated power
in a reverberation chamber (RC) [2]. In addition, USRPs have been
widely used in test beds for experimental evaluation of different com-
munication protocols [3], algorithms [4], [5], network architectures
[6], and a massive MIMO test bed [7]. However, the feasibility of
using USRPs as measurement instruments for passive OTA charac-
terization of antennas is not found in the open literature. In addition,
there cannot be found any comparison with measurements obtained
with standard RF measurement equipment such as a high-end vec-
tor network analyzer (VNA). Here, we aim at filling this gap. The
Manuscript received March 06, 2015; revised November 23, 2015; accepted
December 29, 2015. Date of publication January 14, 2016; date of cur-
rent version March 01, 2016. This work has been supported in part by
Swedish Governmental Agency for Innovation Systems (VINNOVA) within
the VINN Excellence Center Chase, and in part by Swedish Strategic Research
Foundation (SSF) within the Chalmers Microwave Antenna Systems Research
Center CHARMANT.
A. Hussain was with Chalmers University of Technology, SE-41296
Gothenburg, Sweden. He is now with Samsung, Suwon 443-742, South Korea
(e-mail: ah.hussain@samsung.com).
A. A. Glazunov is with the Department of Signals and Systems,
Chalmers University of Technology, SE-41296 Gothenburg, Sweden (e-mail:
andres.glazunov@chalmers.se).
B. Þ. Einarsson was with Chalmers University of Technology, SE-41296
Gothenburg, Sweden. He is now with Merkjafélagið, Reykjavik, Iceland
(e-mail: bjarni@merkjafelagid.is)
P.-S. Kildal is with the Department of Signals and Systems, Chalmers
University of Technology, SE-41296 Gothenburg, Sweden (e-mail:
per-simon.kildal@chalmers.se).
Color versions of one or more of the figures in this communication are
available online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2016.2518211
0018-926X © 2016 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.