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5046 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 11, NOVEMBER 2012
Electronically Steerable 1-D Fabry-Perot Leaky-Wave
Antenna Employing a Tunable High Impedance
Surface
Raúl Guzmán-Quirós, Student Member, IEEE, José Luis Gómez-Tornero, Member, IEEE,
Andrew R. Weily, Member, IEEE, and Y. Jay Guo, Senior Member, IEEE
Abstract—A novel fixed-frequency electronically-steerable
one-dimensional (1-D) leaky-wave antenna is presented. The
antenna is based on a parallel-plate waveguide loaded with a
planar partially reflective surface and a tunable high impedance
surface (HIS), which creates a 1-D Fabry-Perot leaky-waveguide.
The tunable HIS consists of printed patches loaded with varactor
diodes that allow the electronic tuning of the cavity resonance
condition. Using a simple Transverse Equivalent Network, it is
theoretically shown how the variation of the varactors’ junction
capacitance allows the scanning of the antenna pointing angle
from broadside towards the endfire direction at a fixed frequency.
Experimental results of an antenna prototype operating at 5.6
GHz are reported, demonstrating that the new reconfigurable
leaky-wave antenna can provide electronic beam scanning in an
angular range from 9 to 30
Index Terms—Electromagnetic band gap structures, electronic
beam scanning, frequency selective surfaces, high impedances sur-
faces, leaky-wave antennas (LWA), partially reflective surfaces, re-
configurable antennas.
I. INTRODUCTION
SCANNING of the main beam direction is an inherent
property in one-dimensional Leaky-Wave Antennas (1-D
LWAs), which is usually achieved by sweeping the frequency
of the input microwave signal [1]–[3]. For many applications,
however, operation at a fixed frequency is required [4]. Various
technologies and topologies have been proposed in recent
decades to create fixed-frequency electronically-steerable
1-D LWAs [5]–[24]. In all cases, an active device is em-
ployed to produce the electronically controlled change in the
leaky-line boundary conditions, thus altering the leaky mode
(LM) complex propagation constant and pro-
ducing the associated change in its pointing direction (noting
Manuscript received December 22, 2011; revised May 02, 2012; accepted
June 18, 2012. Date of publication August 02, 2012; date of current version
October 26, 2012. This work was supported in part by the Spanish National
project TEC2010-21520-C04-04, in part by the Spanish Regional Seneca
project 08833/PI/08, and in part by the European FEDER.
R. Guzmán-Quirós and J. L. Gómez-Tornero are with the Department
of Communication and Information Technologies, Universidad Politéc-
nica de Cartagena, Cartagena 30202 Spain (e-mail: raul.guzman@upct.es;
josel.gomez@upct.es).
A. Weily and Y. Jay Guo are with the CSIRO ICT Centre, Epping, NSW
1710, Australia.
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2012.2208089
,where is measured from the broadside
direction) [2]. For instance, p-i-n diodes have been added to
dielectric rod LWAs [5], [10] or microstrip LWAs [19], ob-
taining a discrete change in . Similar discrete changes are
reported when introducing photosensitive switches in dielectric
slab LWAs [9], or electronic switches in multi-port microstrip
LWAs [20], [21]. Continuous scan of the main beam has been
obtained in corrugated ferrite LWAs [6], [7], microstrip LWAs
over a ferroelectric substrate [17], photoinduced plasma grating
LWAs [8], and microstrip LWAs loaded with photosensitive
silicon substrates [14], by respectively using magnetic/electric
fields or optical signals to control . However, the most
used active device to electronically control 1-D LWAs in the
microwave range is the varactor diode, which has been applied
to multitude of leaky lines as the slotline [11], the coplanar
waveguide [12], the first higher-order mode microstrip line
[13], the composite right-left handed microstrip line [15], [16],
the microstrip log-periodic line [18], the half-mode microstrip
line [22], [24], and the half-mode substrate-integrated wave-
guide [23].
In this paper, we present a new configuration of reconfig-
urable 1-D LWA to achieve fixed-frequency beam steering,
which is based on a 1-D Fabry-Perot (FP) leaky-waveguide
recently proposed in [25]. In the antenna, varactor diodes are
added to the high impedance surface (HIS) substrate in order
to achieve electronic control of the pointing angle. A simple
transverse equivalent network (TEN) is used to analyze the
LWA and provide a theoretical foundation for the proposed
mechanism to steer the beam. An experimental prototype of the
antenna operating at 5.6 GHz was designed and tested. Good
agreement between theoretical predictions and experimental
results are obtained.
The paper is organized as follows. The working principle of
the new reconfigurable 1-D LWA is described in Section II. It
is shown that, using the TEN, the dispersion of the associated
LM is a function of the varactor junction capacitance. The dis-
persion curves prove essential for the computer-aided design of
the antenna, and particularly for the optimization of the scanning
range. All the TEN analysis results are validated with full-wave
simulations. Section III reports experimental results obtained
from a fabricated prototype operating at 5.6 GHz, showing an
electronically-controlled scanning region from to
as the varactors’ biasing DC voltage is varied
0018-926X/$31.00 © 2012 IEEE
GUZMÁN-QUIRÓS et al.: ELECTRONICALLY STEERABLE 1-D FABRY-PEROT LEAKY-WAVE ANTENNA EMPLOYING A TUNABLE HIS 5047
Fig. 1. (a) Scheme of the proposed reconfigurable 1-D LWA (b) Detail of tun-
able high impedance surface unit cell. ( mm, mm,
mm, mm, mm, mm,
mm, mm, mm,
mm, mm, mm).
from 4.5 V to 18.2 V. Practical limitations of the proposed an-
tenna for scanning angles close to endfire are also discussed. To
finish, Section IV summarizes the conclusions.
II. RECONFIGURABLE 1-D LEAKY-WAV E ANTENNA
The proposed configuration of 1-D reconfigurable LWA and
its main geometrical parameters are shown in Fig. 1(a). The
structure is inspired by the passive antenna presented in [25],
where a parallel-plate waveguide (PPW) was loaded with two
passive printed-circuit boards (PCBs): a top partially reflecting
surface (PRS) and a bottom high impedance surface (HIS) (see
Fig. 1(a)). Both PCBs are formed by periodic arrays of reso-
nant metallic patches printed on thin dielectric substrates. The
two PCBs are separated at a distance , comprising a one-di-
mensional Fabry-Perot resonant cavity which allows the prop-
agation of a perturbed TE leaky mode. Since these types of
antennas combine a host PPW with PCBs, they are known as
hybrid waveguide printed-circuit LWAs [25], [26]. As demon-
strated in [25], this 1-D FP PRS-HIS LWA allows the leakage
rate to be adjusted by changing the resonant length of the PRS
patches ( in Fig. 1), while the resonant length of the HIS
patches ( in Fig. 1) controls the pointing angle at a fixed
design frequency. Using these previous results, a reconfigurable
version of this 1-D LWA is envisaged by introducing varactor
diodes in the middle of the HIS resonant patches, as illustrated
in Fig. 1(b). Similar types of tunable HIS have been used to
design reconfigurable reflectors [27], [28], reconfigurable re-
flectarrays [29], or tunable Frequency Selective Surfaces [30].
Also, an electronically tunable HIS was applied in [31] to trans-
form a surface-wave into a backward or forward leaky-wave,
thus achieving beam scanning from a two-dimensional textured
surface. In [32], [33], a tunable HIS was applied to tune the
operating frequency of low profile 2D FP LWAs, and similar
reconfigurability was demonstrated in [34] for a tunable HIS
fed by a bow-tie antenna. To the best of the authors’ knowl-
edge, it is the first time that a tunable HIS is applied to the
design of a fixed-frequency electronically-scanned one-dimen-
sional Fabry-Perot LWA.
In the following subsection, a simple but accurate TEN is
developed to obtain the complex propagation constant of the
Fig. 2. (a) Cross section of the reconfigurable 1-D FP-PRS-tunable HIS LWA
(b) Transverse Equivalent Network (TEN) of the structure.
leaky-mode which operates in the proposed reconfigurable
LWA, as a function of its main parameters. These dispersion
curves give physical insight into the operating principle and
design rules of the proposed LWA, and lay a theoretical foun-
dation for the steerability of the antenna main beam.
A. Analysis of Reconfigurable LWA Using a TEN
The Transverse Resonance Method (TRM) is a simple and
powerful analysis tool widely used in the design of LWAs [2],
[35], [36]. Particularly, as it was demonstrated in [25], the TRM
allows efficient analysis and design of the passive version of
the 1-D FP PRS-HIS LWA. For the analysis of the reconfig-
urable version of this LWA, it is necessary to extend the TRM
approach proposed in [25]. The key feature of the TRM is the
development of an appropriate Transverse Equivalent Network
(TEN) that accurately models the cross section of the antenna,
which in our case is shown in Fig. 2. The constituent parts of this
TEN are well described in [25], with the most important parts
being the equivalent admittances which model the top PRS and
the bottom HIS [ and in Fig. 2(b)]. In this paper, the
necessary novelty required to model the tunable HIS admittance
as a function of the varactors’ junction capacitance is intro-
duced in the TEN formulation.
As explained in [37], the pole-zero method proposed by Maci
et al. in [38] can be expanded to take into account more op-
timization parameters beyond frequency in the design of 1-D
LWA. For instance, the effect of the resonating length of the PRS
patches [ in Fig. 1(a)] can be efficiently modeled by using
the following analytical pole-zero expansion for [37]:
(1)
where the location of the poles and the zeros depend on the
longitudinal wavenumber , which is related to the leaky-mode
incidence (or radiating) angle by .
Fig. 3 illustrates the magnitude and phase of the reflection
coefficient presented by the PRS [ in Fig. 2(b)] as a func-
tion of for the design frequency of 5.6 GHz, and for three
different incidence angles. As explained in [25], must be
properly chosen to tune the PRS reflectivity in order to control
5048 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 11, NOVEMBER 2012
Fig. 3. Reflection coefficient of the PRS (magnitude and phase) seen by a plane
wave for different incidence angles as a function of at 5.6 GHz.
the antenna radiation efficiency. The rest of the geometrical pa-
rameters of the PRS are fixed, so patches of width
mm separated at a distance mm, are printed on
FR4 substrate ( mm). As shown
in Fig. 3, good agreement is observed between the analytical
pole-zero model of the PRS admittance (1) and full-wave re-
sults obtained with commercial Finite Element Method (FEM)
solver HFSS [49]. mm is chosen to provide a PRS
reflectivity above 0.9 for all scanning angles, as shown in Fig. 3.
The main novelty of the proposed TEN is in the modeling of the
tunable HIS. In this case, the following pole-zero expression is
employed for , which is now dependent on the varactors’
junction capacitance
(2)
In the antenna design, we use a one dimensional version of the
tunable HIS configuration proposed in [29] and [32], so that a
row of metallic patches of width mm, length
mm, separated at a distance mm, are printed on
grounded Rogers RO4230 substrate (
mm). Each HIS patch is divided inthex-directionbytwoequal
patches separated by a gap ( mm) where two varactor
diodes are placed as illustrated in Fig. 1(b).
As was demonstrated in [25], the boundary condition pre-
sented by the HIS changes drastically as the physical resonating
length is varied, thus resulting in a direct mechanism to
control the TE leaky-mode cut-off frequency and therefore
the pointing angle at a fixed frequency. Yet, this control could
only be mechanically achieved in [25], as the whole PCB should
be replaced in order to vary . The proposed reconfigurable
LWA overcomes this disadvantage, since the effective resonant
length of the HIS patches can be varied by properly modifying
the varactors’ junction capacitance [29]. This is shown in
Fig. 4(a), where the reflection phase presented by the tunable
HIS [ in Fig. 2(b)] provides PMC condition
at different frequencies (5.15 GHz, 5.65 GHz and 6.15 GHz) as
is varied (0.35 pF, 0.23 pF, and 0.15 pF). These results were
obtained using the pole-zero analytical expression of as a
function of frequency, for different values of and .When
the frequency is fixed to 5.6 GHz, the behavior of
Fig. 4. Reflection phase of the tunable HIS seen by an incident plane wave for
different incidence angles (a) as a function of frequency for different values of
(b) as a function of at 5.6 GHz.
can be analytically modeled using (2), obtaining the HIS re-
flection phase as a function of for different shown in
Fig. 4(b). Very good agreement with FEM results is observed for
all the range of (from 0 pF to 0.35 pF) and for all incidence
angles. The HIS becomes a grounded dielectric slab for
pF , changing to PMC sheet for
pF, and to PEC sheet for
pF. According to the results reported in [25], increasing cor-
responds to a larger , thus showing a direct relation be-
tween and the effective resonant length of the HIS patches.
Since can be controlled by the varactors’ reverse polariza-
tion voltage [27]–[34], electronic control of the pointing angle
at a fixed design frequency should be possible by combining the
1-D FP leaky cavity and the tunable HIS. To theoretically con-
firm this fact, the behavior of the radiating leaky mode must be
studied as a function of .
B. Effect of in the Leaky-Mode Dispersion Curves
Obtaining the leaky-mode dispersion curves as a function
of is essential for the electronic beam steering application
presented. The unknown leaky-mode complex wavenumber
can be obtained from the TEN shown in Fig. 2,
by solving the corresponding Transverse Resonance Equation
(TRE) [25], [37]
(3)
Fig. 5 shows the frequency dispersion for the normalized
phase and leakage constants as is varied
from 0.1 pF to 0.23 pF. As expected, the TE leaky-mode
GUZMÁN-QUIRÓS et al.: ELECTRONICALLY STEERABLE 1-D FABRY-PEROT LEAKY-WAVE ANTENNA EMPLOYING A TUNABLE HIS 5049
Fig. 5. Leaky-mode dispersion curves as a function of (mm)
(a)–(b) Dispersion with frequency (c) Dispersion with at 5.6 GHz.
cutoff frequency decreases as increases. This results in a
continuous rise of the leaky-mode phase constant at a fixed fre-
quency as is increased, as it is shown in Fig. 5(c) for 5.6
GHz. Again, very good agreement is obtained between TRM
and full-wave FEM results, validating the proposed TEN. As
it can be seen in Fig. 5(c), is varied from values close to
zero when pF, to values close to one when
pF. Since the leaky-mode pointing angle is given by
, it is expected that fixed-frequency beam-scanning from
broadside to endfire can be realized by controlling in the
aforementioned range [0.01 pF, 0.23 pF]. It must be noticed that
at this stage we use a lossless model of the tunable HIS, not
taking into account on the dispersion analysis the effects of the
varactor series resistance and the substrates losses.
To give more physical insight into the working principle of
the proposed reconfigurable LWA, Fig. 6 illustrates the TE
leaky-mode electric field distribution in the cross section of the
FP cavity for different values of at the design frequency of
5.6 GHz. For low values of the HIS behaves as a grounded
Fig. 6. Leaky-mode electric-field pattern inside the FP PRS-tunable HIS cavity
(obtained from FEM) at 5.6 GHz for different values of .
Fig. 7. Computed H-plane normalized radiation patterns for the proposed re-
configurable LWA at 5.6 GHz for different (antenna length ).
slab as shown in Fig. 4(b) and the leaky-mode is resonating in
the metallic cavity of height , providing maximum horizontal
field at , as it can be seen in Fig. 6 for pF.
However, as is increased to 0.23 pF, the HIS tends to behave
as a PMC sheet which induces maximum field at its interface,
as shown in Fig. 6. The boundary conditions seen by the leaky
mode resonating in the FP cavity are strongly changed as
is varied. As a result, the leaky-mode transverse wavelength
is modified from when pF to
when pF, as illustrated in Fig. 6. This enlargement in
the transverse wavelength implies the subsequent reduction
in the longitudinal wavelength , and therefore a rise in the
leaky-mode longitudinal phase constant as is
increased, as predicted by the results shown in Fig. 5(c).
Once the leaky-mode complex propagation constant has been
obtained as a function of , the associated H-plane radiation
pattern can be directly obtained [2], as shown in Fig. 7 for
the case of a LWA of length at 5.6 GHz. As ex-
pected, the scanning angle is swept from nearly broad-
side towards the endfire direction as is increased, following
the curve plotted in Fig. 5(c). Particularly, Fig. 7 shows
the results for pF
pF ,and
pF . The beam direction and the
beamwidth predicted from the TRM are in very good agreement
with full-wave FEM simulations performed in a 3D model of the
whole reconfigurable LWA. A strong reflected lobe appears for
pF, due to the very low leakage-rate associated with
high values of [see Fig. 5(c)], which results in poor radia-
tion efficiency and a large amount of energy being reflected at
thefar-endoftheLWA.Thisfactlimits the maximum scanning
5050 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 11, NOVEMBER 2012
Fig. 8. Leaky-mode scanning-angle as a function of for different values of
the Fabry-Perot cavity height, ,at5.6GHz.
angle of the proposed reconfigurable LWA to 40 , as it will be
explained in detail in Section III.
C. Optimization of the Fabry-Perot Cavity Height
The physical height of the FP cavity is a key geometrical
parameter which determines the scanning range and sensitivity
of the proposed reconfigurable LWA. All previous results were
computed using the optimum value mm. As
can be seen in Fig. 8, a wider scanning range is obtained in this
case, so that the cavity height makes the leaky-mode cut-off con-
dition to be coincident with the minimum value of
. In this situation, the dynamic range of the varactors is fully
used and the minimum scanning angle is close to broadside. On
the other hand, as discussed in [39], the maximum achievable
pointing angle in this type of 1-D LWA is produced when the
HIS presents a PMC resonance (which in our case corresponds
to the value pF).
Above the HIS PMC resonance, a sudden fall of the leakage
rate and a rapid divergence of the phase constant as shown
in Fig. 5(c) limit the scanning range to higher angles. Fig. 8
shows that for the scanning angle can be swept from
for pF to for pF,
as previously described. However, if ,theminimum
value of does not correspond to the leaky-mode cut-off
point but to a higher scanning state. As a result,
the minimum scanning angle is located further from broadside
at pF in Fig. 8 for mm). Since
the maximum pointing angle given by the HIS PMC resonance
(pF) is , the scanning range is reduced
to [30 ,40 ] instead of [5 ,40 ]. Also, it can be observed that
the sensitivity of with is lower for this case, due to the
nonlinear response of the varactor.
On the other hand, for shorter FP cavities ( , shown
in dotted red line in Fig. 8) the leaky mode is below cutoff for
most of the range of variation of . As it can be seen in Fig. 8
for mm, the leaky modes below cutoff from
pF to pF, missing this region of the varactors’ dy-
namic range. As a result, the scanning starts at pF
with , but then it is very soon limited to a maximum
angle of at pF. This reduces the scan-
ning range and increases the sensitivity. Fig. 8 shows that the
Fig. 9. Photographs of (a) Manufactured reconfigurable LWA prototype (b)
tunable HIS phase agile cell (c) Radiation pattern measurement experimental
set-up and (d) S parameters measurements set-up.
optimum cavity provides the most linear dependence between
and and the highest scanning range. Once more, ex-
cellent agreement is found in the validation of the TRM disper-
sion curves and FEM results.
III. EXPERIMENTAL RESULTS
The following section describes the experimental results car-
ried out on a prototype fabricated at CSIRO ICT Centre, shown
in Fig. 9. The antenna is designed to operate at the fixed fre-
quency of 5.6 GHz, and the radiating aperture has a length of
. As illustrated in Fig. 9(a), three constituent parts
in the reconfigurable LWA can easily be distinguished: a par-
allel-plate waveguide, a passive PRS printed-circuit, and a tun-
able HIS. To excite the TE leaky-mode of the tunable 1-D
FP cavity, a horizontal coaxial probe is used as proposed in
GUZMÁN-QUIRÓS et al.: ELECTRONICALLY STEERABLE 1-D FABRY-PEROT LEAKY-WAVE ANTENNA EMPLOYING A TUNABLE HIS 5051
Fig. 10. Measured S parameters vs frequency for different (a) (b) .
[25]. The dimensions of this feeding probe were optimized to
obtain maximum input matching at the operating midpoint of
pF, which results in better performance across most
of the dynamic range of .
A detailed photograph of the phase agile cell used for the
tunable HIS, including the biasing network proposed in [32],
is depicted in Fig. 9(b). Metelics’ MGV125-08 varactor diodes
(0805 package) are used for the tunable HIS, providing a range
in from 0.055 pF to 0.6 pF as reverse bias voltage is
tunedfrom20V to2V . The varactor’s tolerance for
is pF according to the manufacturer datasheet.
The measured parameters in the 5 GHz–6 GHz frequency
range are presented in Fig. 10 for different values of bias voltage
. As expected, the matching band is shifted to lower frequen-
cies as is decreased (and is increased), in accordance with
the expected theoretical shift in the leaky-mode frequency dis-
persion presented in Fig. 5. These results are also consistent with
data reported in previous reconfigurable LWAs using varactors
[27]–[34]. At the fixed operation frequency of 5.6 GHz, Fig. 11
shows the simulated and measured S parameters as a function
of . As expected, the matching and the transmission
coefficients are affected as is varied. Simulated
data predict optimum operation with dB and
dB for V(which corresponds to
pF, according to the datasheet).
As is varied from this value of 7.17 V ,theTE leaky-
mode field distribution is strongly perturbed (as it was theoreti-
cally illustrated in Fig. 6), thus decreasing the matching between
Fig. 11. Measured S parameters vs at 5.6 GHz.
the coaxial probe and the FP cavity. It is important to note that,
as previously mentioned, the coaxial probe dimensions were op-
timized for the aforementioned operating point pF
(V). As a result, poorer matching is obtained at
other operation points of the varactor’s dynamic range. In par-
ticular, the mismatch increases up to dB for
V ( pF) and to dB for
V ( pF). Experimental data report a similar ten-
dency for the measured S parameters as a function of , ob-
taining dB and dB at the optimum
tuning point V ( pF) and similar dete-
rioration for other values of . As can be seen, measured data
is shifted to lower values with respect to simulations. This
shift is attributed to the varactors’ tolerance errors, as demon-
strated next.
The electronic control over the LWA radiation pattern is
shown in Fig. 12. Fig. 12(a) depicts the measured normalized
radiation pattern at 5.6 GHz for different values of the applied
bias voltage . As it can be seen, the main beam is scanned
from for V ( pF) to
for V ( pF). Figs. 12(b)
and (c) show the agreement between theory and experiments. In
particular, Fig. 12(b) plots with red crosses the measured scan-
ning response ( vs of the proposed reconfigurable
LWA, confirming the continuous electronic scanning from
34.2 to 9.2 as is increased from 4.5 V to 18.2 V .
The theoretical leaky-mode scanning response extracted from
the TRM is plotted with a continuous blue line, while FEM
simulations are represented with blue circles. Very good agree-
ment is observed between these three set of results. However,
the measured scanning response is slightly shifted with respect
to TRM and FEM results, showing discrepancies in that
are below . This can be attributed to the aforementioned
pF varactors’ tolerances in . To demonstrate this fact,
error curves computed by applying a shift in of pF
and pF are represented with shaded zones in Fig. 12(b).
As can be seen, the measured scanning response lies inside the
pF region, well below the diodes’ tolerance (
pF).
Fig. 12(c) compares the theoretical and measured leaky-mode
radiation patterns for three illustrative values of covering
the entire scanning range. As it was demonstrated in Fig. 12(a),
5052 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 11, NOVEMBER 2012
Fig. 12. Radiation response of reconfigurable LWA at 5.6 GHz vs (a) Mea-
sured radiation patterns (b) Pointing angle scanning (c) Comparison between
measured and theoretical (TRM) radiation patterns.
the main beam direction presents higher discrepancies between
theory and experiments for scanning angles close to endfire
(lower values). Despite this fact, the 3 dB beamwidth and the
secondary lobes distribution are in accordance with leaky-mode
far-field patterns [2], thus confirming the leaky-wave radiation
mechanism for the proposed reconfigurable antenna. Table I
summarizes the pointing angle obtained for three opera-
tion points inside the scanning region. As it is shown, maximum
error is obtained at lower voltages that correspond to larger scan
angles ( V , ). It is worthy to note
that the scanning response predicted by the lossless TRM is in
agreement with FEM and experiments, despite that the latter
take into account the losses created by the varactors series re-
sistance and materials. This confirmsthatthermallossesdonot
have a strong influence on the leaky-mode phase constant, and
the proposed lossless TRM is useful for the design of this an-
tenna.
TAB L E I
BIAS VOLTAGE AND POINTING ANGLE FOR THREE OPERATING POINTS OF THE
RECONFIGURABLE 1-D LWA
Fig. 13. Results at 5.6 GHz vs (a) Measured gain, directivity and total effi-
ciency, (b) Estimated efficiencies and (c) Simulated reflection coefficient (mag-
nitude and phase) of lossy HIS.
Finally, Fig. 13 summarizes the experimental performance of
the reconfigurable LWA in terms of gain and efficiency. Fig.
13(a) shows the variation of the gain as a function of ,re-
porting maximum measured gain of dBi for
V ( pF, ), while simulations pre-
dict maximum dBi for V (
pF, ). Moreover, as is decreased from this op-
timum point, a significant drop in the gain is observed. The min-
imum value of gain measured is dBi for
GUZMÁN-QUIRÓS et al.: ELECTRONICALLY STEERABLE 1-D FABRY-PEROT LEAKY-WAVE ANTENNA EMPLOYING A TUNABLE HIS 5053
V ( pF), which corresponds to an angle close to
.
Similar behavior is obtained with simulations, thus making
it difficult to obtain large scan angles. As it was previously ex-
plained, this fact is caused by higher mismatch (see Fig. 11) and
by the fall of the leakage rate (see Fig. 5(c)) that take place at
high values of (lower ), due to the associated PMC res-
onance of the HIS for high (see Fig. 6). However, the
LWA shows a stable gain above dBi for V ,
which corresponds to scanning angles below .
As also plotted in Fig. 13(a), this gain translates into a total
efficiency %, reaching %
for – V . Measured and
simulated total efficiency in Fig. 13(a) are in concor-
dance, observing % in the range – V
, whereas tends to 3% for
V. To understand the gain andefficiency re-
sponse of the LWA, Fig. 13(b) separates into its different
constitutive efficiencies [41], [42]
(4)
where the mismatch efficiency is computed from the measured
input reflections [41], the leaky-mode ra-
diation efficiency is estimated from the theoretical leakage rate
[42], and the ohmic efficiency has
been attributed to two contributions: one part due to the dielec-
tric losses and another term due to dissipation in the varactors
series resistance: [41]. Since the separated
ohmic efficiencies (due to dielectrics and varactors) are diffi-
cult to compute from experiments [41], they have been esti-
mated from full-wave simulation data. Equation (4) gives, as
afirst order approximation, a good insight into the origin of
the gain drop for large scan angles. Particularly, it can be seen
in Fig. 13(b) that the mismatch efficiency is quite high
(over 85%) for the entire dynamic range. On the other hand, the
leaky-mode radiation efficiency strongly decreases from
100% at Vto 52% at V
, being consistent with the drop of the leakage
rate predicted in Fig. 5(c) for high values of (i.e., high values
of ). The dielectric efficiency results predict a fall from
%at ( V )to %
at ( V ), which is caused by the electric
field concentration at the HIS substrate for the PMC regime (see
Fig. 6 for pF). Due to the same high concentration
of the modal fields, an increase in the current density flowing
across the diodes occurs at this PMC operational point, raising
the heat dissipation at their series resistance. As a result, the var-
actors’ ohmic efficiency strongly drops for scanning an-
gles tending to endfire, observing a decrease from %
at ( V )to %at
( V ).
This increase of thermal losses in the HIS as the antenna
scans is demonstrated in Fig. 13(c). The reflection coefficient
of the HIS is simulated with HFSS for each operating point
of the designed antenna. As it can be observed, the
HIS absorption increases as the LWA scans towards endfire, and
for there is an abrupt drop to .This
is related with the strong change of the HIS phase, which oper-
ates close to the PMC resonance for ,
in accordance with the lossless HIS response shown in Fig. 4(b)
for pF. This result agrees with other published works
that report significant increase in the HIS ohmic losses when op-
erating close to the PMC resonance, and which has even been
widely applied to create planar microwave absorbers [44]–[46].
Therefore, it can be concluded that the gain drop for large scan
angles is unavoidable and it is due to two main reasons: the de-
crease in the leaky-mode radiation efficiency, and the increase
in the varactors resistance losses. This latter effect could have
also been predicted by the proposed TEN model, by introducing
the varactors’ series resistance in the tunable HIS equivalent ad-
mittance (2), and the losses tangent in the substrate equivalent
transmission line sections, as done in [44]–[46].
IV. CONCLUSION
A new type of electronically scannable one-dimensional
LWA, which is based on a tunable Fabry-Perot (FP) cavity
formed by a parallel-plate waveguide (PPW), a partially re-
flective surface (PRS), and a varactor-loaded tunable high
impedance surface (HIS), has been presented in the paper. The
proposed reconfigurable LWA topology allows to electroni-
cally modify the resonant condition of the TE leaky mode
inside the FP cavity, resulting in a variation of the pointing
direction at a fixed frequency as the varactors’ bias voltage is
changed. Leaky-mode dispersion curves have been obtained
from a simple but accurate Transverse Equivalent Network,
which rigorously takes into account the effect of the varactors’
junction capacitance in the phase response of the tunable HIS.
These modal curves are very useful for the design of the main
parameters of the antenna, in order to optimize the beam-scan-
ning range. Leaky-mode results are in very good agreement
with full-wave simulations and experiments, which have shown
a continuous scanning from 9 to 30 in a prototype operating
at the fixed frequency of 5.6 GHz, as the varactors’ bias voltage
is varied from 18 V to 5 V . The scanning range is limited
in the endfire region due to the drop in the leakage rate and
the increase of thermal losses, which are associated with the
perfect magnetic conductor (PMC) condition of the tunable
high impedance surface. An interesting advantage of this new
reconfigurable topology resides in its potential application to
two-dimensional FP LWAs [32], [33], [35], [40] and to new
metasurface LWAs [47], [48], so that the electronic scanning
could be extended for both elevation and azimuthal planes.
This type of electronically steerable LWA presents a low-cost
potential alternative to more expensive phased-arrays [48].
ACKNOWLEDGMENT
The authors would like to thank C. Holmesby for fabricating
the waveguide, R. Shaw for soldering the surface mount compo-
nents, and M. García-Vigueras for her insightful and interesting
comments.
5054 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 11, NOVEMBER 2012
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GUZMÁN-QUIRÓS et al.: ELECTRONICALLY STEERABLE 1-D FABRY-PEROT LEAKY-WAVE ANTENNA EMPLOYING A TUNABLE HIS 5055
Raúl Guzmán Quirós (S’12) was born in Cartagena
(Murcia), Spain, in 1986. He received the Telecom-
munications Engineer degree from Universidad
Politécnica de Cartagena (UPCT), Cartagena, Spain,
in 2009, where he is currently working towards the
Ph.D. degree.
In 2010, he joined the Department of Communi-
cation and Information Technologies, UPCT, where
he is involved as a Research Assistant in a regional
project related to RFID indoor location systems. His
current research interests include reconfigurable an-
tennas, analysis and design of novel active leaky-wave antennas and location
systems.
José Luis Gómez Tornero (M’06) was born in
Murcia, Spain, in 1977. He received the Telecom-
munications Engineer degree from the Polytechnic
University of Valencia (UPV), Valencia, Spain, in
2001, and the “laurea cum laude” Ph.D. degree in
telecommunication engineering from the Technical
University of Cartagena (UPCT), Cartagena, Spain,
in 2005.
In 1999 he joined the Radio Communications De-
partment, UPV, as a research student, where he was
involved in the development of analytica l and numer-
ical tools for the automated design of microwavefilters in wav eguide technology
for space applications. In 2000, he joined the Radio Frequency Division, In-
dustry Alcatel Espacio, Madrid, Spain, where he was involved with the devel-
opment of microwave active circuits for telemetry, tracking and control (TTC)
transponders for space applications. In 2001, he joined the Technical University
of Cartagena (UPCT), Spain, as an Assistant Professor. From October 2005 to
February 2009, he held de position of Vice Dean for Students and Lectures af-
fairs in the Telecommunication Engineering Faculty at the UPCT. Since 2008,
he has been an Associate Professor at the Department of Communication and In-
formation Technologies, UPCT. His current research interests include analysis
and design of leaky-wave antennas and the development of numerical methods
for the analysis of novel passive radiating structures in planar and waveguide
technologies.
Dr. Gómez Tornero received the national award from the foundation EPSON-
Ibérica to the best Ph.D. project in the field of Technology of Information and
Communications (TIC) in July 2004. In June 2006, he received the Vodafone
foundation-COIT/AEIT (Colegio Oficial de Ingenieros de Telecomunicación)
award to the best Spanish Ph.D. thesis in the area of Advanced Mobile Commu-
nications Technologies. This thesis was also awarded in December 2006 as the
best thesis in the area of Electrical Engineering, by the Technical University of
Cartagena. In February 2010, he was appointed CSIRO Distinguished Visiting
Scientist by the CSIRO ICT Centre, Sydney.
Andrew R. Weily (S’96-M’01) received the B.E. de-
gree in electrical engineering from the University of
New South Wales, Australia, in 1995, and the Ph.D.
degree in electrical engineering from the University
of Technology Sydney (UTS), Australia, in 2001.
From 2000 to 2001 he was a research assistant
at UTS. He was a Macquarie University Research
Fellow then an ARC Linkage Postdoctoral Research
Fellow from 2001 to 2006 with the Department of
Electronics, Macquarie University, Sydney, NSW,
Australia. In October 2006 he joined the Wireless
Technology Laboratory at CSIRO ICT Centre, Sydney. His research interests
are in the areas of reconfigurable antennas, EBG antennas and waveguide
components, leaky wave antennas, frequency selective surfaces, dielectric
resonator filters, and numerical methods in electromagnetics.
Y. Jay Guo (SM’96) received a Bachelor Degree
and a Master Degree from Xidian University in
1982 and 1984, respectively, and a Ph.D. Degree
from Xian Jiaotong University in 1987, all in China.
In 1997, he was awarded a Ph.D. degree by the
University of Bradford, U.K., for his research
achievements in Fresnel antennas. His research
interest includes reconfigurable antennas and radio
systems, antenna arrays, wireless positioning and
multi-gigabit wireless communications.
He has been with the Commonwealth Scientific
and Industrial Organization (CSIRO) since 2005, managing a number of port-
folios of research programs including Smart and Secure Infrastructure, Broad-
band Networks and Services, Broadband for Australia and Safeguarding Aus-
tralia. From August 2005 to January 2010, he served as the Research Director
of the Wireless Technologies Laboratory in CSIRO ICT Centre. Prior to joining
CSIRO, he held various senior positions in Fujitsu, Siemens and NEC in the
U.K.
Dr. Guo has served in the organizing and technical committees of numerous
national and international conferences. He is the Patronage and Publicity Chair
of IEEE ICC2014, TPC Chair of 2010 IEEE Wireless Communications and Net-
working Conference (WCNC), and TPC Chair of 2007 and 2012 IEEE Interna-
tional Symposium on Communications and Information Technologies (ISCIT).
He has been the Executive Chair of Australia China ICT Summit since 2009. He
was a Guest Editor of the special issue on “Antennas and Propagation Aspects
of 60–90 GHz Wireless Communications,”IEEET
RANSACTIONS ON ANTENNAS
AND PROPAGATION, and Special Issue on “Communications Challenges and Dy-
namics for Unmanned Autonomous Vehicles,”IEEEJ
OURNAL ON SELECTED
AREAS IN COMMUNICATIONS (JSAC). He is the recipient of 2007 Australian
Engineering Excellence Award, 2007 and 2012 CSIRO Chairman’s Medal and
2012 CSIRO Newton Turner Award. He has published three technical books
“Fresnel Zone Antennas,” “Advances in Mobile Radio Access Networks”and
“Ground-Based Wireless Positioning,” 80 journal papers and 130 refereed inter-
national conference papers. He holds 18 patents in wireless technologies. He is
an Adjunct Professor at University of New South Wales, Macquarie University,
University of Canberra, all in Australia, and a Guest Professor at the Chinese
Academy of Science (CAS) and Shanghai Jiaotong University. He is a Fellow
of IET.
