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Abstract—In this paper, we propose an approach to realize
substrate integrated waveguide (SIW)-based leaky-wave antennas
(LWA) supporting continuous beam scanning from backward to
forward above the cutoff frequency. First, through phase delay
analysis, it was found that SIWs with straight transverse slots
support backward and forward radiation of the –1 order mode
with an open-stopband in between. Subsequently, by introducing
additional longitudinal slots as parallel components, the
open-stopband can be suppressed, leading to continuous beam
scanning at least from −40° through broadside to 35°. The
proposed method only requires a planar structure and obtains less
dispersive beam scanning compared with a composite right/left-
handed LWA. Both simulations and measurements verify the intended
beam scanning operation while verifying the underlying theory.
Index Terms—Leaky-wave antenna, backward radiation,
broadside radiation, forward radiation, open-stopband.
I. INTRODUCTION
LEAKY-WAVE antenna (LWA) refers to a type of
traveling-wave structure based on a transmission line
loaded with radiating elements [1]. They were first designed
using a rectangular waveguide with slots [2]. The arrangement
of the slots determines the polarization and the radiation
direction of the LWA [3]. Periodic loading of the same
components on each unit cell of the LWA leads to a high
This work was supported in part by the Open Project Program of the State
Key Laboratory of Millimeter Wave under Grants K201403. Corresponding
author: F.-Y. Meng (e-mail: blade@hit.edu.cn).
Y.-L. Lyu, and P.-Y. Wang are with the Department of Microwave
Engineering, Harbin Institute of Technology, Harbin, 150001 China (e-mail:
lvyuelonglvyuelong@126.com, and pywang@hit.edu.cn).
X.-X. Liu is with Beijing Aerospace Microsystems Institute, Beijing,
100094, China (e-mail: liuxiaoxin__0014pm@163.com)
D. Erni is with the Faculty of Engineering, Laboratory for General and
Theoretical Electrical Engineering (ATE) and the CENIDE – Center for
Nanointegration Duisburg–Essen, University of Duisburg–Essen, D-47048
Duisburg, Germany (e-mail: daniel.erni@uni-due.de).
F.-Y. Meng and Q. Wu, and are with the Department of Microwave
Engineering, Harbin Institute of Technology, Harbin 150001, China, and also
with the State Key Laboratory of Millimeter Waves, Nanjing 210096, China
(e-mail: blade@hit.edu.cn and qwu@hit.edu.cn ).
C. Wang and N.-Y. Kim is with the Department of Wireless Communication
Engineering, Kwangwoon University, Seoul 139-701, Korea
(kevin_wang@kw.ac.kr and e-mail: nykim@kw.ac.kr).
directivity, and a frequency-dependent beam scanning ability.
These unique characteristics makes the LWA attractive in many
applications such as e.g. frequency modulated continuous wave
radar [4], real-time spectrum analysis [5], and field pattern
synthesis [6].
The beam scanning range is one of the most concerned issues
in LWA design. Early LWAs radiate electromagnetic (EM)
power only forward (towards end-fire) as frequency goes up
[7-9], or both forward and backward (i.e. backfire) but not
broadside, which is due to the open-stopband (OSB) effect [10,
11]. In [12], the OSB is attributed to the unmatched unit cell
structure of LWAs. At the frequency of broadside radiation, the
reflected waves of each unit cell are in phase, and hence no EM
power is fed into the LWA for broadside radiation. To obtain
seamless beam scanning from backward to forward, a few types
of microstrip LWAs have been proposed to suppress the OSB by
means of reflection cancellation [12], impedance matching [12,
13], and implementing asymmetric unit cells [14, 15]. On the
other hand, the continuous beam scanning of LWA can also be
achieved by making use of the concept of balanced composite
right/left-handed (CRLH) transmission line [16]. In the past
decade, varieties of CRLH-based LWAs were designed
utilizing microstrip lines [17], mushroom structures [18, 19],
coplanar striplines [20], coplanar waveguides [21, 22],
rectangular waveguides [23-25], and substrate integrated
waveguides (SIW) [26-28]. Among all these waveguide
topologies rectangular waveguide and SIW-based LWAs have
demonstrated their advantages such as low loss and high power
capacity. However, most of the SIW or rectangular
waveguide-based CRLH LWAs operate from below the cutoff
frequency [4, 24, 27]. This is because cutoff rectangular
waveguides and SIWs possess inherent shunt inductances,
which is constitutive for realizing CRLH transmission lines.
However, such CRLH LWA is usually hard to match due to the
highly dispersive nature of the cutoff waveguide [24, 27].
CRLH LWAs above the cutoff frequency have been proposed
in [25] using a rectangular waveguide with inductive irises, but
this LWA is bulky and relies on complex 3D structures, which
are hard to fabricate.
Very recently, some SIW-based LWAs have been designed
to operate above the cut-off frequency such as the LWA with
long slots [29, 30], and the forward radiating LWA with
Leaky-Wave Antennas Based on Non-Cutoff
Substrate Integrated Waveguide Supporting
Beam Scanning from Backward to Forward
Yue-Long Lyu, Student Member, IEEE, Xiao-Xin Liu, Peng-Yuan Wang, Daniel Erni, Member, IEEE,
Qun Wu, Senior Member, IEEE, Cong Wang, Senior Member IEEE, Nam-Young Kim, Member IEEE,
and Fan-Yi Meng, Senior Member, IEEE
A
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transverse slots as presented in [31-33]. However, the LWAs in
[29, 30] require ridges to achieve backward, broadside, and
forward radiation, relying on multilayer substrates for
fabrication purposes. Furthermore, the LWAs in [31-33] are not
able to realize broadside radiation due to the OSB as will be
discussed later in this paper. Although the OSB can be
suppressed in LWAs consisting of reflection-cancelling
transverse slot pairs [34, 35], this type of LWAs usually radiate
linearly polarized EM waves with their polarization directions
along the longitudinal axis of the SIW. Radiation of other
polarizations requires adaption of the transverse slot pair. For
example, an oblique slot is added between the transverse slot
pair to achieve 45° linear polarization [36]. In this paper, we
propose a LWA design method to suppress the OSB by
utilizing an impedance-matched LWA unit cell. The required
unit cell consists of both transverse and longitudinal slots on the
top layer of the SIW as the series and shunt loading elements,
respectively. By careful design, a series of LWAs are realized
capable of EM wave emission with a seamless transition
through broadside. The proposed LWAs are easy to fabricate
owing to its single layer structure. In addition, the proposed
LWAs in this paper radiate linear polarized EM waves with
polarization directions perpendicular to the longitudinal axis of
the SIW. However, they are also promising for circularly
polarized radiation as shown later. A prototype of the proposed
LWAs is fabricated and experimentally validated as proof of
concept where the measured results agree perfectly well with
the simulated data.
Fig. 1. The structure of the SIW-based LWA with transverse slots, where (a) is
the realized model, (b) is a rectangular waveguide model for simulation
purposes, (c) is the equivalent circuit of the unit cell, and (d) is the LWA
alignment in the coordinates.
The remainder of this paper is organized as follows: Section
II provides a discussion on SIW LWAs with transverse slots,
aiming at providing a theoretical foundation of the underlying
radiation properties, whereas Section III gives the theoretical
description, the model design, and optimization of the new
planar structure with both transverse and longitudinal slots for
continuous beam scanning through broadside. Section IV
presents the measurement of a fabricated prototype, and finally,
a conclusion is drawn in Section V.
II. SIW-BASED LWA WITH TRANSVERSE SLOTS
The structure of the SIW-based LWA with transverse slots is
sketched in Fig. 1(a), which is a similar structure as discussed in
[31]. The substrate is covered with copper on both sides
forming a waveguide, together with two rows of metallic vias
along the wave propagation direction. The thickness and the
width of the SIW are h and w, respectively. The diameter of the
metallic vias is d, and the period of the vias is s. For the
following theoretical analysis including numerical simulations
conducted with the CST MWS software package, the SIW is
simplified to a flattened dielectric-filled rectangular waveguide
with an effective width weff as obtained from [37] to reduce both
the numerical complexity and simulation time [as shown in Fig.
1(b)]. In this simplified case, for the TE10 mode, the phase
constant β can be written as following, where the relative
permittivity of the substrate is εr = 3.66.
2
2
0reff
kw
(1)
The transverse slots on the top of the SIW are implemented
to perturb the current distribution, not only for radiating EM
power to the free space, but also for introducing an additional
phase delay in the underlying transmission line. The width,
length, and period of the slots are ws, ls, and p, respectively.
The unit cell of the LWA is modeled with two transmission line
segments and a series loaded impedance in between as sketched
in Fig. 1(c). Its S parameters can be described as
, (2)
where Z0 is the port impedance, and Z represents the impedance
corresponding to the transverse slot. The phase increment
caused by the transverse slot is arg(Z/Z0+2). The effective
phase constant βeff and the attenuation constant αeff of the unit
cell are obtained from [21]
1
1
1Re cos 2
1Im cos 2
eff
eff
AD
p
AD
p
, (3)
where A and D are the elements of the ABCD matrix of the unit
cell, which can be calculated through (2) using the classic
conversion formulas [38]. For the alignment of the LWA in the
coordinate system shown in Fig. 1(d), the radiation direction of
SIW LWA can be derived as
0
00
0
00
/2
2 / 2 /
/
2
2 / 2 /
j p j p
unit j p j p
ZZ ee
Z Z Z Z
ZZ
ee
Z Z Z Z
S
3
0
arcsin eff
rad k
, (4)
where k0 is the wavenumber of free space. When |βeff| < k0, EM
waves are within the fast wave regime and thus have the potential
to be radiated from the LWA, otherwise the EM waves propagate
as (guided) slow wave modes and cannot be radiated at all. The
positive, zero, and negative value of the θrad indicate forward,
broadside, and backward radiation, respectively.
Considering a parametrized SIW with transverse slots accor-
ding to TABLE I, the impedance characteristics of the slot
should be investigated first, having Z approximately described
as
1tan 2 /2
2e eff
Z j Z f ls R
, (5)
where Z0, εeff, and R are the characteristic impedance, effective
permittivity, and the radiation resistance of the slot,
respectively. The resonant frequency of the slot is denoted as
0/2 eff
f c ls
, (6)
where c is the speed of light. Z is then extracted from the
simulated S-parameters of the unit cell of the SIW LWA and its
frequency response is depicted in Fig. 2 when normalized to the
port impedance Z0 of the waveguide. As shown in Fig. 2, Z has
an increasing positive imaginary part within a large frequency
band up to f0 = 18.83 GHz, which means that the transverse slot
remains inductive within this frequency range. The inset of Fig.
2 depicts the surface current distribution of the unit cell at 11.8
GHz. The phenomenon that the surface current is strong at the
ends of the slot but weak at its center verifies the inductive
property of the transverse slot at this frequency. On the other
hand, the real part of Z [i.e. R in (5)] is relatively small except at
20.25 GHz. The phase increment arg(Z/Z0+2) caused by the
transverse slot is displayed in Fig. 3. The calculated k0∙p and β∙p,
and the simulated effective phase βeff ∙ p and the attenuation αeff ∙
p of one unit cell are also shown in this figure.
TABLE I
STRUCTURE PARAMETERS OF THE SIW-BASED LWA WITH TRANSVERSE
SLOTS
Parameters
w
d
s
weff
h
ws
ls
p
Values
(mm)
10
0.8
1.6
10
0.762
0.45
6
15
Fig. 2. The simulated impedance Z of the transverse slot normalized to the port
impedance.
In Fig. 3, one can observe three distinct radiation bands
where |βeff ∙ p| < k0p and αeff ∙ p is small. The corresponding main
lobe direction of a LWA constructed from cascading 10 unit
cells is depicted in Fig. 4. The lower boundaries of these
radiation bands are determined by the Bragg reflection
condition β∙p = mπ [16], where m = 0, 1, and 2, respectively.
Both arg(Z/Z0+2) and β∙p in concert increase with the frequency
and leads to an increasing βeff ∙ p too. For the n = 0 harmonic, βeff
∙ p rises from 0 to π in the band from the cutoff frequency of 7.9
GHz to 9.1 GHz, resulting in forward beam scanning from 10°
to 70° [see in Fig. 4] as discussed in [33]. As the frequency
continuously increases, the n = −1 harmonic is able to leak EM
waves. The decreasing |βeff ∙ p| from 9.8 GHz to 11.8 GHz and
the increasing |βeff ∙ p| from 13.4 GHz to 15 GHz (as depicted in
Fig. 3) indicates a backward (from −60° to broadside) and a
forward beam scanning (from broadside to 40°), respectively.
The frequency band from 11.8 GHz to 13.4 GHz is the so-called
OSB, where βeff ∙ p = 0. According to (1), as S11 of the unit cell
cannot be zero due to the mismatched structure, all the reflected
waves of each unit cell superimpose in phase in the OSB at
which the EM power is barely fed into the SIW structure, which
makes the αeff ∙ p considerably large. Therefore, seamless beam
scanning through broadside cannot be achieved due to the
presence of the OSB.
Fig. 3. The comparison of calculated k0∙p, simulated |βeff ∙ p| and αeff ∙ p of the
unit cell, simulated arg(Z/Z0+2), and the calculated β∙p.
Fig. 4. The simulated main lobe direction of the LWA consisting of 10 unit cells
of different radiation modes.
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III. SIW-BASED LWA WITH CONTINUOUS BEAM SCANNING
FROM BACKWARD TO FORWARD
A. Theory Description
As discussed in the last section, SIW-based LWAs with one
transverse slot in the unit cell are confronted with the
unavoidable emergence of an OSB caused by the severe return
loss. The OSB suppression can be achieved by using a reflecti-
on-cancelling unit cell structure, i.e. a radiation element pair
with a separation distance of 1/4 λg [12, 34]. The OSB can be
also closed by using a transformer in the unit cell for proper
impedance tuning [12]. In this section, we try to suppress the
OSB by achieving an impedance-matched unit cell structure.
As the transverse slot loaded on the SIW only introduces the
series impedance Z, a shunt admittance Y cascaded with Z should
be added in the equivalent circuit of the unit cell to accomplish
impedance matching, as shown in Fig. 5(a). Based on microwave
network theory, the S parameters of the equivalent circuit
displayed in Fig. 5(a) can be calculated as
11 21
21 11
00
11 00
21 00
/
2/
2
2/
unit
jp
jp
SS
SS
Z Z YZ ZY
Se
Z Z YZ ZY
Se
Z Z YZ ZY
S
. (7)
According to (7), to achieve impedance matching, i.e. S11 = 0,
Z and Y should follow the relationship described as
00
Z
YZ Z Z
. (8)
Furthermore, the real and imaginary parts of the shunt
admittance Y are expressed as
2
0 0 0 0
20
0
22
00
Re Re Im
1
Re 1
Im Re
Im 1
Re Im
1
Z Z Z
YAZ Z Z Z
ZZ
YZ
AZ
ZZ
AZZ
. (9)
It’s worth noting that Re[Z]/Z0 ≪ 1 in the frequency band far
below the resonant frequency of the slot [see in Fig. 2], and
hence both of the real and imaginary parts of Y are positive in
the band of interest, which renders Y capacitive.
The periodic loading with relatively long longitudinal slots
on top of the broad wall of the waveguide can be an effective
approach to accomplish parallel capacitive loading in the
corresponding equivalent circuit of the unit cell. As longitudinal
slots disturb the transverse current of the waveguide, they
function as a parallel component, and when properly adjusting
its length its resonant frequency drops below the frequency of
interest, as expected for capacitive loading. The combination of
the proper longitudinal and transverse slots is promising to
approximately realize the structure shown in Fig. 5(a).
B. Initial Design
An initial design of the unit cell for a SIW with both
transverse and longitudinal slots is illustrated in Fig. 5(b). The
longitudinal and transverse slots are aligned like a symmetrical
“T” with a distance ds. The length and width of the longitudinal
slot are l_lon, and w_lon, respectively. The design process
consists of two steps: first at the desired frequency for
broadside radiation, i.e. the transition frequency, the period of
the unit cell should be determined by β∙p = 2π to ensure that
adjacent unit cells radiate EM power in phase; second the
parameters of the slots including their sizes and relative
position should be properly adjusted to avoid an OSB.
Fig. 5. The structure of LWA with both longitudinal and transverse slots,
where (a) shows the equivalent circuit of the unit cell, (b) is the initial design,
and (c) sketches the modified design.
Fig. 6. The simulated surface current distribution on the surface of (a) the
initial design and (b) the modified design, with E-field probes marked.
It should be noted that the longitudinal slots are much longer
than the transverse slots and hence prone to EM wave radiation,
5
leading to a polarization direction of the proposed LWA in this
section that is perpendicular to that of the LWA with only
transverse slots. Therefore, and for the sake of a linear
polarized LWA, the length of the transverse slots should be
reduced to suppress its contribution to the radiation field (cross
polarization).
Comprehensive numerical simulations are conducted to
verify all theoretical predictions. All parameter values of the
simulation results are listed in TABLE II. For better physical
understanding the mechanism of the proposed unit cell structure,
the surface current distribution on the top layer of the unit cell of
the initial design is simulated and illustrated in Fig. 6(a). It can be
seen that the current distribution around the transverse slot is
similar to what is shown in the inset of Fig. 2: it is strong at the
slot ends, but weak at the slot center, meaning that the transverse
slot is inductive. However, the surface current around the
longitudinal slot is strong at the center, but weak at the slot ends.
This reversal phenomenon compared to the case of the transverse
slot indicates that the longitudinal slot achieves capacitive
loading. Therefore, this unit cell agrees well with the
requirements of the equivalent circuit shown in Fig. 5(a).
TABLE II
STRUCTURE PARAMETERS OF THE SIW-BASED LWA WITH TRANSVERSE AND
LONGITUDINAL SLOTS
Parameter
Initial Design
(mm)
Modified Design
(mm)
Prototype (mm)
w
-
-
10.5
d
-
-
0.8
s
-
-
1.6
weff
10
10
10
h
0.762
0.762
0.762
ws
0.45
0.45
0.45
Ls
3.5
3.5
4
p
15
7.5
7.5
l_lon
12.2
10
9.5
w_lon
0.45
0.45
0.45
ds
2.5
1
1
N
10
19
19
Fig. 7. Simulated dispersion diagram of the unit cell of the LWA with both
transverse and longitudinal slots.
Both phase delay |βeff∙p| and attenuation αeff∙p of the unit cell
of the initial design are depicted in Fig. 7. The decrease and
increase of |βeff∙p| with frequency indicate the backward and
forward radiation, respectively. The attenuation of the unit cell
is negligible from 9 to 13.8 GHz. The simulated S parameters of
the LWA retrieved from the corresponding N cascaded unit
cells are depicted in Fig. 8. One can observe that in the total
passband, a S11 magnitude peak occurs at 11.8 GHz due to the
weak OSB effect between the backward radiation band and the
forward radiation band. This is because perfect parameter
optimization is hard to accomplish (9) for full elimination of the
OSB. In fact, the OSB barely affects the broadside radiation as
this peak is much lower than −10 dB.
Fig. 8. Simulated spectral response of the magnitude of S parameters for both
the initial design and the modified design.
The main lobe direction and the associated gain of the initial
design are illustrated in Fig. 9. As shown, the main lobe direction
of this LWA scans from –60° at 9 GHz to 40° at 14 GHz (under-
going large gain fluctuations), with a maximal realized gain of
13.8 dB at 20° (13.2 GHz). Fig. 10(a) plots the main lobe patterns
in the y-o-z plane at three typical frequencies. The narrow beam
scans smoothly through broadside at the transition frequency of
11.8 GHz, which is consistent with what is shown in Fig. 9. The
smooth and steady beam scanning with respect to frequency can
be understood by analyzing the phase delay of the unit cell. The
scattering parameter S21 of the unit cell under condition (9) can be
rewritten as
. (10)
Fig. 9. Main lobe direction and realized gain of both the initial design and
modified design as a function of frequency.
As deduced from (10), the phase delay of one unit cell is only
21 jp
Se
6
determined by the phase constant β of the host waveguide
because the transverse inductive slot and the longitudinal
capacitive slot counteract each other and, therefore do not affect
the phase delay of the unit cell. Compared to CRLH LWAs that
rely on a cutoff waveguide with a highly dispersive impedance,
the proposed LWA makes use of the SIW above the cutoff
frequency, resulting in a much steadier frequency behavior.
In summary, the initial design in this subsection features
continuous beam scanning from backward to forward with
smooth transition through broadside. The OSB is well suppressed
and low return loss is obtained. Although some of its radiation
properties need to be improved, such as the low leakage rate in
the backward radiation band [see in Fig. 9], and the relatively
high cross polarization level (varying from −15 dB to −20 dB)
as shown in Fig. 10(a), the modified design proposed in the next
subsection presents a potential solution to these deficiencies.
(a)
(b)
Fig. 10. Simulated frequency-dependent scanning of the main lobe patterns of
(a) the initial LWA design, and (b) the modified design.
C. Modified Design
The modified structure is designed as shown in Fig. 5(c). In
the modified unit cell, the longitudinal slots are loaded alternately
on opposite sides of the transverse slots. For broadside radiation,
the phase delay of each unit cell should be reduced to π for the
adjacent longitudinal slot to obtain in-phase radiation of the
emitted EM power. As a result, the period of the unit cell is
reduced by half, yielding radiation contributions from this
modified LWA structure that are more densely distributed
along the propagation direction (i.e. the y axis) compared to the
initial design. The radiation rate per unit length of the total
LWA is therefore improved. As the period of the modified unit
cell becomes smaller than the length of the longitudinal slots,
the coupling between the adjacent unit cells becomes stronger
than in the initial design, and thus some parameter adjustment
for the modified LWA is needed to suppress a potentially
emerging OSB. The parameters of the optimized structure are
listed in TABLE II, and the simulated surface current
distribution on the top layer of the unit cell is illustrated in Fig.
6(b), which is similar to the surface current of the initial design.
The alternately aligned longitudinal slots give rise to a balanced
current distribution with respect to the transverse axis of the
SIW on the top layer of the unit cell, which leads to a more
symmetrical radiation pattern in the x-o-z plane compared to the
initial design.
(a)
(b)
Fig. 11. Simulated spectral response of (a) the normalized magnitude, and (b)
the phase of the E-field at the probe set in the initial design.
The simulated magnitudes of S parameters as a function of
frequency are depicted in Fig. 8. The passband of the modified
design approximately overlaps that of the initial design but with
broader bandwidth. The peak magnitude of S11 in the passband
caused by the weak OSB effect appears at the transition
frequency of 11.8 GHz. In the modified design the OSB
suppression of is harder to achieve compared to the initial
design as the |S11| reaches larger values in the passband. As
shown in Fig. 8, it is interesting that there is a slight increase of
both the |S21| and the |S11| at 11.8 GHz in the modified design,
7
which was also be observed in [23, 26, 39], where dips in the
spectral response of |S21| are more common at transition
frequency [4, 25, 26] (which is also true for the initial design).
The opposite frequency behavior of the S21 magnitude of both,
the initial design and the modified design reveal the complex
effect of the OSB on the reflected power, radiated power, and the
remainder power.
The frequency dependence of the simulated main lobe
direction and the associated antenna gain of the modified design
are depicted in Fig. 9. In comparison to the results of the initial
design, the modified LWA has nearly the same beam scanning
range as that of the initial design. This modified LWA reaches
its maximum gain of 14.89 dB at 13.7 GHz with a radiation
beam pointing at 22.7°. With nearly the same total length, the
realized gain of the modified LWA is much higher and its
frequency dependence is remarkably steadier compared to the
initial design. The main lobe radiation patterns of the modified
LWA at three typical operation frequencies are shown in Fig.
10(b). Continuous beam scanning from backward to forward
can be observed. In addition, the resulting cross polarization
level of the modified design is extremely low [better than −40
dB as shown in Fig. 10(b)], much lower than in the initial
design. To shed light on this phenomenon, four E-field probes
are set at the center of the adjacent slots of both the initial
design and the modified design as illustrated in Fig. 6.
(a)
(b)
Fig. 12. Simulated spectral response of (a) the normalized magnitude, and (b)
the phase of the E-field at the probe set in the modified design.
Fig. 11 depicts the normalized magnitude and phase of the
E-field at the probes set in the initial design. Probe A and C
sense the E-field of the adjacent longitudinal slots, while Probe
B and D sense the E-field of the adjacent transverse slots. As
expected, the E-field at the longitudinal slots is much stronger
than at the transverse slots, making the co-polarization direction
perpendicular to the longitudinal axis of the SIW as already
described in subsection B. The phase differences of both the
adjacent longitudinal slots and the adjacent transverse slots are
2π (i.e. in phase) at the transition frequency of 11.8 GHz. For
the case of the modified design, the normalized magnitude and
phase of the E-field at the probes are plotted in Fig. 12. As
shown in Fig. 12(a), similar to the initial design, the E-field at the
longitudinal slots of the modified design is much stronger than
that at the transverse slots. However, the phase differences of the
E-field of the adjacent transverse slots is only π at the transition
frequency as shown in Fig. 12(b), which makes the E-field
excited by the adjacent transverse slots superpose anti phase, but
leads to the in-phase superpose of the radiated power of the
longitudinal slots as the alternately alignment of the longitudinal
slots introduces an additional phase increment of π between the
adjacent longitudinal slots. As a result, the cross-polarization
level of the modified design is much lower than that of the initial
design.
It is interesting to observe that the E-field phase difference
between the longitudinal and transverse slots is approximately
π/2 at the transition frequency in both the initial design and the
modified design. This phenomenon is similar to the case
reported in [15]. Therefore, the unit cell composed of both
transverse and longitudinal slots is capable to radiate circularly
polarized EM wave once the radiated powers of transverse slot
and longitudinal slot are balanced. Therefore, the control of the
polarization of the proposed SIW-based LWA is more flexible
than prior LWA designs constructed from the
reflection-cancelling units.
IV. EXPERIMENTAL VERIFICATION
For verification purposes, we fabricated a prototype of the
modified design in Section III as shown in Fig. 13. This
prototype is fed by SMA connectors through microstrip-to-SIW
transitions [40] as the prototype unit cell is well designed to
allow adequate matching to the characteristic impedance of the
host SIW. The prototype is fabricated using Rogers 4350 (εr =
3.66) substrates. The structure parameters are listed in TABLE
II.
Fig. 13. Realized prototype of the SIW-based LWA.
The S-parameter measurements are conducted using an
Agilent N5227A microwave vector network analyzer (VNA),
and the results are depicted in Fig. 14, together with the
simulation results (from CST MWS). Both simulation and
measurement of the magnitude of S11 displays similar shapes
8
and tendencies in the spectral response. Due to additional
reflections caused by the SMA connectors and potential
production variations, the measured |S11| spectrum presents
shallower resonance dips compared to the simulated spectrum.
Hence, the measured magnitude of the S21 prototype’s spectral
response is also slightly deviating from the corresponding
simulated S21 spectrum due to the worsened S11.
Fig. 14. The measured and simulated magnitude of S parameters of the
prototype as a function of frequency.
Fig. 15. The measured and simulated radiation patterns of the prototype.
Fig. 16. The main lobe direction and realized gain of the prototype (both
simulation and measurement).
The measured and simulated normalized radiation patterns of
the antenna prototype are depicted in Fig. 15, showing a
virtually perfect agreement. The measured and simulated main
lobe direction and associated gain are depicted in Fig. 16. The
well agreed measured and simulated results demonstrate that the
main radiation beam of the prototype scans at least from −40° at
9 GHz to 35° at 14 GHz with smooth transition through
broadside at 11.4 GHz. The maximal realized gain of the
prototype is 12 dB.
V. CONCLUSION
In this paper, we provide a design method for LWAs based
on non-cutoff SIW supporting continuous beam scanning from
backward to forward. First, the radiation mechanisms of the
SIW with transverse slots are thoroughly analyzed. Then, with
the additional introduction of longitudinal slots, the performance
of the original LWA with transverse slots has been significantly
improved yielding impedance matching together with the
elimination of the OSB between the −1 order backward and
forward radiation. This allows a scanning range of the radiation
pattern from –40° to 35° with a fractional bandwidth of 55.3%.
The proposed LWA operates above cutoff frequency, which is
thus easy to feed, and features a smoother beam scanning with
regard to frequency compared to a CRLH LWAs that rely on
cutoff waveguides. Both simulation and experiment underpin the
underlying theory and renders the proposed LWA design very
promising for efficient beam steering in ultra-compact mm wave
antenna applications.
ACKNOWLEDGEMENT
The authors thank the reviewers and editors for their
constructive comments and suggestions, which helped to
improve the quality of this paper.
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Yue-Long Lyu (S’14) received the B.E., M.E.
degrees in Microwave Engineering from Harbin
Institute of Technology (HIT), Harbin, China, in
2012 and 2014, respectively. Now he is working
toward the Ph.D. degree in the department of
microwave engineering, HIT. His current research
interests are beam steering antenna, tunable
microwave device, and wireless power transfer.
He was a recipient of the student travel award
from IEEE international conference on microwave
magnetics (ICMM), July, 2014, Sendai, Japan and
the student paper contest award (honorable prize) from the 3rd IEEE
Asia-Pacific conference on antennas and propagation (APCAP), July, 2014,
Harbin, China. He was also awarded as the outstanding graduate of HIT in 2012,
and 2014, respectively.
Xiao-Xin Liu was born in Tangshan, China, 1991.
He received the B.S. and M.S. degrees in
electromagnetic field and microwave technology
from Harbin Institute of Technology, Harbin, China,
in 2013 and 2015, respectively. He is currently
working at Beijing Aerospace Microsystems
Institute, Beijing, China. His research interests
include the design and measurement of leaky-wave
antennas, the design and application of frequency
selective surfaces and RF circuits for broad-band
wireless communications.
10
Peng-Yuan Wang received the B.E. degree in
communication and information systems from
Dalian Maritime University (DLMU), Dalian,
China, in 2014. Now he is working toward the M.E.
degree in the department of microwave engineering
of Harbin Institute of Technology (HIT). His
current research interests are tunable microwave
devices and MIMO system with single RF chain.
He was awarded as the outstanding graduate of
Dalian in 2014.
Daniel Erni (S’88–M’93) received the Diploma
degree in electrical engineering from the
University of Applied Sciences (HSR),
Rapperswil, Switzerland, in 1986, the Diploma
degree in electrical engineering and Ph.D. degree
from ETH Zürich, Zürich, Switzerland, in 1990
and 1996, respectively. Since 1990, he has been
with the Laboratory for Electromagnetic Fields
and Microwave Electronics, ETH Zürich. He
was the founder, and from 1995 to 2006, the
Head of the Communication Photonics Group,
ETH Zürich. Since October 2006, he has been a
Full Professor of general and theoretical electrical engineering at the University
of Duisburg–Essen, Duisburg, Germany. His current research includes
advanced data transmission schemes (i.e. O-MIMO) in board-level optical
interconnects, optical on-chip interconnects, ultra-dense integrated optics,
nanophotonics, plasmonics, quantum optics, and optical and electromagnetic
metamaterials. The latter with a distinct emphasis on biomedical engineering,
namely for advanced RF excitation schemes in magnetic resonance imaging.
On the system level, he has pioneered the introduction of numerical structural
optimization into dense integrated optics device design. Further research
interests include science and technology studies as well as the history and
philosophy of science with a distinct focus on the epistemology in engineering
sciences. He is a member of the Editorial Board of the Journal of
Computational and Theoretical Nanoscience.
Dr. Erni is a Fellow of the Electromagnetics Academy. He is a member of the
Center for Nanointegration Duisburg–Essen (CeNIDE). He is also as a member
of and the Applied Computational Electromagnetics Society (ACES), the Swiss
Physical Society (SPS), the German Physical Society (DPG), and the Optical
Society of America (OSA). He is an associated member of the Swiss
Electromagnetics Research Centre (SEREC).
Qun Wu (M’93–SM’05) received his B.Sc.
degree in radio engineering, the M.Eng. degree in
electromagnetic fields and microwaves, and the
Ph.D. degree in communication and information
systems from Harbin Institute of Technology
(HIT), Harbin, China, in 1977, 1988, and 1999,
respectively.
Since 1990, he has been with School of
Electronics and Information Engineering, HIT,
China, where he is currently a Professor and the
head of Department of Microwave Engineering.
He is also a director of Center for Microwaves and
EMC. He worked as a Visiting Professor with
Seoul National University (SNU), Seoul, Korea, from 1998 to 1999, and
Pohang University of Science and Technology, from 1999 to 2000. He was a
visiting Professor with National University of Singapore, from 2003 to 2010.
He published several books and over 100 international and regional refereed
journal papers. His recent research interests mainly include electromagnetic
compatibility, metamaterials, and antennas.
Prof. Wu was a recipient of the Science and Technology Award from
Heilongjiang Provincial Government in 2010. He is a Member of Microwave
Society of the Chinese Institute of Electronics. He is a technical reviewer for
several international journals. He is also a vice chair of IEEE Harbin section,
and chair of IEEE Harbin EMC/AP/MTT joint society chapter. He worked as a
chair or member in the TPC of international conferences for many times. He is
also invited to give a keynote report or invited papers in some international
conferences for many times.
Cong Wang (S’08–M’11–SM’16) was born in
Qingdao, Shandong Province, China in 1982. He
received the B.S. degree in Automation Engineering
from Qingdao Technological University (China) in
2005, the M.S. and Ph. D. degrees in Electronic
Engineering from Kwangwoon University (Korea) in
2008 and 2011, respectively. He is currently working
at the same university as an assistant professor. In
2007, he was a research engineer at Mission
Technology Co., Ltd R&D Department,
Gyeonggi-do, Korea, where he was involved in the
development of microwave components and devices using Hybrid and LTCC
technology. He joined NanoENS Co., Ltd R&D Department that same year. His
work includes passive device modeling, passive device design, and fabrication
process development and optimization. He has also co-authored two books and
published more than 190 papers in domestic and international journals and
conferences. He also has over 40 patents registered in Korea. His major
interests include RFIC/MMIC design and semiconductor fabrication
development such as GaAs integrated passive device, silicon-based LED
module fabrication and packaging, AlGaN/GaN HEMT, and various kinds of
smart sensors and their applications which are emerging technologies of today.
Nam-Young Kim (M’10) received the Masters
and Ph.D. degrees in electronic engineering from
the State University of New York (SUNY) at
Buffalo, Buffalo, NY, USA, in 1991 and 1994,
respectively, the Masters and Ph.D. degree in
theology from Midwest University, St. Louis, MO,
USA, in 2004 and 2006, respectively. He was then
a Research Scientist with the Center for Electronic
and Electrooptic Materials (CEEM), SUNY at
Buffalo. In 1994, he joined the Department of
Electronic Engineering, Kwangwoon University,
Seoul, Korea, as a Professor. His main research
focus is RF integrated circuits (RFICs), RF nano-devices, and RF nano-bio
devices. He is the founder of the RFIC Research Center and also serves as
Director of the Fusion Technology Center, Kwangwoon University. He leads
the RFIC and compound semiconductor related research group at Kwangwoon
University. He has authored or coauthored 175 refereed journal papers, 28
books, and 343 refereed conference papers. He holds over 116 patents and
semiconductor design patents.
Fan-Yi Meng (S’07–M’09-SM’15) received the
B.S., M.S., and Ph.D. degrees in electromagnetics
from the Harbin Institute of Technology, Harbin,
China in 2002, 2004, and 2007, respectively. Since
August 2007, he has been with the Department of
Microwave Engineering, Harbin Institute of
Technology, where he is currently a Professor. He
has coauthored four books, 40 international
refereed journal papers, over 20 regional refereed
journal papers, and 20 international conference
papers. His current research interests include
antennas, electromagnetic and optical
metamaterials, plasmonics, and electromagnetic compatibility (EMC).
Dr. Meng was a recipient of several awards including the 2013 Top Young
Innovative Talents of Harbin Institute of Technology, the 2013 CST University
Publication Award, the 2010 Award of Science and Technology from the
Heilongjiang Province Government of China, the 2010 “Microsoft Cup” IEEE
China Student Paper Contest Award, two Best Paper Awards from the National
Conference on Microwave and Millimeter Wave in China (2009 and 2007,
respectively), the 2008 University Excellent Teacher Award of the National
University of Singapore, the 2007 Excellent Graduate Award of Heilongjiang
Province of China, and the Outstanding Doctor Degree Dissertation Award of
the Harbin Institute of Technology.
