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Transactions on Antennas and Propagation
1
60 GHz Compact Larger Beam Scanning Range
PCB Leaky-wave Antenna using HMSIW for
Millimeter-Wave Applications
Anirban Sarkar, Member, IEEE and Sungjoon Lim, Member, IEEE
Abstract—A half-mode substrate integrated waveguide, based
printed circuit board (PCB) technology, is employed to construct
a novel V-band leaky-wave antenna incorporating composite
right/-left handed (CRLH) media. Including modified half-
sinusoidally varied inter-digital capacitive slots on the radiating
edge of the HMSIW enables the feature of effective CRLH
media to produce adequate radiation coverage and bandwidth
with improved side-lobe level and higher gain. Unit cell char-
acteristics and space harmonic analysis were thoroughly inves-
tigated through dispersion and Bloch impedance analysis. By
appropriately tuning the physical parameters of the design, the
balanced condition was obtained at broadside frequency of 60
GHz. The highly directed radiated fan beam covered a scanning
range of 120◦with beam steering from -72◦to 48◦of the
visible space. The proposed antenna operated over 55-65 GHz
with maximum realized gain of 14.5 dBi, maximum beamwidth
of 9.15◦, better than 15 dB side-lobe level and 20 dB cross-
polar radiation. All simulated responses showed reasonably good
accordance with experimental data thereby validating the novel
design of the proposed leaky-wave structure for millimeter-wave
wireless applications.
Index Terms—Frequency beam scanning, Half-mode substrate
integrated waveguide, Composite right/-left handed, leaky-wave
antenna.
I. INT ROD UC TI ON
MILLIMETER-WAVE (mm-wave) frequencies in the
electromagnetic wave spectrum have gained much at-
tention due to their applications in several areas such as
compact radar systems for navigation [1], target detection
and tracking [2], high-definition video streaming [2], wireless
back-haul links [3], high resolution imaging radars and mining
[4], collision avoidance [5], [6], 5th generation mobile com-
munications etc. It not only demands novel technological solu-
tions for market requirements but also requires new standards
for advanced antenna designs. For instance, IEEE 802.11ad
wireless networking standard requires significant antenna gain
of around 16 dBi [7]–[8], broad bandwidth of 14.6%(57-
66 GHz) to handle large data traffic, high cross polar level
to avoid interference, large beam scanning range of ±50◦
[7]–[8] with simpler feeding beamforming network, low cost
for bulk production etc. [7]-[9]. Recently, the standardization
regulatory bodies have focused their attention upon unlicensed
V-band especially ETSI: 57-66 GHz [3]. Due to its numerous
benefits such as higher data rate services, frequency reuse to
A. Sarkar and S. Lim are with the School of Electrical and Electronics Eng.
Chung-Ang University, Seoul, Korea (e-mail: anirban.skr227@gmail.com and
sungjoon@cau.ac.kr)
“This research was supported by the National Research Foundation of Korea
(NRF) grant funded by the Korea government (MSIT) (2018R1A4A1023826)”
accommodate dense population demands, broader bandwidth,
small wavelength etc. [10]-[17], makes this band unlicensed or
light licensed. The relevant techniques such as low temperature
co-fired ceramic (LTCC) [18], MEMS bulk micromachining
technology [19] etc. have been adopted to achieve desired
antenna performances at mm-wave frequencies. However, they
suffer from complex feeding network [18], limited radiation
coverage like 10◦[18] and 31◦[19], lower gain like 11.4 dBi
[18] and 11.7 dBi [19], and design complexity due to being
multilayer and high free space path loss for implementation
over higher frequencies [18]. Leaky-wave antennas (LWAs)
which belong to the traveling wave antenna family, are one
of the fittest candidates to overcome this, due to its several
unique advantages including highly directional beams, fre-
quency beam steering capability, adequate bandwidth, fair gain
etc [20], [21]. Several LWAs have been recently reported at
V-band adopting various technologies. In [18], a LTCC based
substrate integrated image guide has been used to design mm-
wave leaky-wave antenna within 58-67 GHz. LTCCs have been
widely used for high frequency applications particularly mm-
wave antenna arrays, but the required multilayered fabrication
makes the overall design somewhat complex. Reference [19]
proposed another V-band LWA (52-63 GHz) based on MEMS
technology. Also, in [22], another new technique has been
adopted to design LWA at 60 GHz where inset dielectric
waveguide has been built using 3D printing technology with
sinusoidally modulated metal grove. Moreover, a metasurface
loaded Luneburg lens fed highly directive LWA has been pro-
posed at V-band [23] but the performance is not satisfactory.
A recent LWA combined with a partially reflecting surface
achieved high gain (22.6 dBi) at 60 GHz, focused on some
medical applications [24].
It is to be noted that the previous V-band LWAs [20]-
[24] have either inadequate scanning range with broader
beamwidth or suffer from multilayered fabrication, enhanced
fabrication cost, bulky, significant design complexity and com-
pactness. Substrate integrated waveguides (SIWs) are one of
the promising solutions which claim the advantages of being
light weight, low cost, low loss, ease of manufacturing and
easy integration with other planar circuits over conventional
metallic waveguides [25]. Consequently, extensive research
has been conducted on SIW based LWAs, e.g. [26], [27].
These wave-guiding structure based LWAs are mostly uniform
or quasi-uniform types that work in the fast wave region
under the fundamental mode. Hence, scanning range for these
kind of antennas, is usually delimited only within the forward
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Transactions on Antennas and Propagation
2
region. Reference [28] proposed a SIW based Ku-band LWA,
providing enhanced scanning range by introducing sinusoidal
modulation of the slots profile. Periodic perturbations can also
be introduced in non-radiating geometry to design periodic
LWAs [29], [30]. Also, several works were reported on half-
mode SIW based leaky-wave structures [31]-[35]. Recently,
composite right/left-handed (CRLH) transmission lines [36]
gained much attention in designing LWAs because such TL
inherently supports backward to forward continuous beam
scanning. CRLH media has also been frequently incorporated
with SIWs, half-mode SIWs (HMSIWs) [37] and eighth-mode
SIWs (EMSIWs) to design high performance LWAs [38]-[44].
However, most of the previous studies were at lower frequen-
cies, whereas, dielectric loss and fabrication tolerances become
severe issues for higher frequencies (e.g. mm-wave bands),
particularly when the array dimension becomes large. Many
technologies have been used to design SIW based antennas and
filters at mm-wave frequency range, including conventional/-
printed circuit boards (PCBs) [45], [46], photoimageable thick-
film technology [47], LTCC [48], etc. PCB based V-band
LWAs using SIW have been recently proposed, achieving
high efficiency and larger radiation coverage [49], [50], but
performance was unsatisfactory. From the above discussion it
can be inferred that it is quite challenging to design a LWA
at mm-wave frequency range having a larger beam steering
range, adequate operating bandwidth maintaining a high gain,
design simplicity, and compactness.
This paper considers these problems in the context of a
novel larger radiation coverage, high gain, simple LWA based
on SIW planar PCB technology. Significantly improved op-
erating bandwidth, narrow beamwidth, and cross-polar levels
(CPL) were achieved by implementing the modified half-
sinusoidally varied inter-digital capacitive slots. This corruga-
tion of the HMSIW radiating edge also ensures the presence of
CRLH media within the proposed geometry, thereby providing
larger beam steering range from backward to forward includ-
ing broadside directions. The proposed antenna performance
was thoroughly optimized using high-frequency structure sim-
ulator (HFSS) and validated experimentally. The proposed
structure can steer the radiated beam from -72◦to +48◦of the
visible space by varying frequency from 55 to 65 GHz with
the radiator length of 5.6λ0. The broadside radiation occurred
at 60 GHz with side lobe level (SLL) better than 15 dB. Also,
the backward to forward beam scanning by the narrow radiated
beam with improved cross-polar level (∼20 dB) provides
good beam isolation for two different frequencies, suitable
for communication in dense traffic environment. The paper is
organized by mentioning the HMSIW based CRLH unit cell
design methodology in Section II, antenna design principle
in Section III, results and analysis in Section IV and finally
drawn a conclusion in Section V.
II. IN IT IA L HAL F-M OD E SIW BAS ED C RL H UN IT CE LL
DES IG N
A. Selection of Proposed Unit Cell
Firstly, the V-band half-mode SIW (HMSIW) transmission
line (TL) was designed and fabricated for the frequency range
Fig. 1. Layout of the proposed V-band LWA with periodicity p= 2.4 mm.
Fig. 2. Performance comparison between the unit cells with uniformly
oriented IDC slots and proposed modified half-sinusoidally varied IDC slots.
(a) Total attenuation constant of the structure, (b) Radiated power, (c)
Corresponding gain variations.
of 50-65 GHz. The tapered microstrip line fed by V-band wave
launcher was used for the excitation purpose. The difference
between simulated and measured insertion loss is obtained
of 1.5 dB with good impedance matching within operating
band. The overall dimension of the fabricated HMSIW section
(including ground plane) are 2.3cm ×0.3cm ×0.038cm.
Further, uniformly oriented inter-digital capacitive slots on the
top of the HMSIW are introduced to provide an extra series
capacitance (or left-handed capacitance, CL) along with pre-
existing shunt inductance (or left-handed inductance, LL) due
to metallic vias. The upper metallic layer and lower ground
plane form a series inductance (or right-handed inductance,
LR) and distributed shunt capacitance (or right-handed ca-
pacitance, CR). Hence, the overall unit cell geometry enables
the CRLH property. The corrugated HMSIW radiating edge
widens the impedance bandwidth and also narrows the antenna
beamwidth. Our primary intention was to develop a structure
having CRLH property and exhibiting higher radiated power
for gain enhancement maintaining compactness. For this
purpose, the uniformly oriented IDC (inter-digital capacitive)
slots based CRLH HMSIW unit cell was designed. As shown
in Fig. 2(a), the total attenuation constant (αt) of the uniformly
oriented and half-sinusoidally varied IDC slots are compared.
Figure 2(b) shows the maximum radiated power along the
structure which is a function of total attenuation constant (αt)
and antenna length (L). Because αtof the half-sinusoidal de-
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Transactions on Antennas and Propagation
3
Fig. 3. Dispersion analysis and its impact on radiation pattern for the unit cell
with (a) uniformly oriented slots and (b) proposed modified half-sinusoidally
varied IDC slots.
Fig. 4. Normalized Bloch impedance characteristics for different types of
corrugation (a) exponential, (b) uniform and (c) proposed.
sign is larger than that of the uniform design, the total radiated
power of the half-sinusoidal design is larger than that of the
uniform design. In addition, as the antenna length increases,
the radiated power is increased and then limited to 60–70%
with prolonged saturation region beyond the frequency range
of 55–65 GHz. Next, the corrugation of the half-sinusoidally
varied IDC slots were placed within HMSIW unit cell. As a
result, not only the radiated power was increased significantly
up to 97%due to larger attenuation constant (Fig. 2(a) and
(b)) resulting into higher gain (Fig. 2(c)), its saturation region
also came forward, making the design compact.
The dispersion diagram for the initially selected uniformly
oriented IDC slot based unit cell is depicted in Fig. 3(a) within
the operating frequency band of 55–65 GHz having smaller
operating region shown by normalized phase constant (|β|/k0)
(red curve). The loss profile in terms of normalized attenuation
constant is also shown by blue curve in Fig. 3(a). From the
figure one can infer that the high value of losses not only
increases the 3dB beamwidth (∼16◦) of the radiation pattern,
also enhances the side lobe level. On contrary, the dispersion
analysis for the proposed unit cell was also performed at the
balanced point 60 GHz covering the LH band from 55-60
GHz and RH band from 60-65 GHz as shown in Fig. 3(b)
(red curve). Bloch impedance ZBwas also calculated from
an infinite periodic structure where the periodic boundary
condition was assigned on a single unit cell in ANSYS
HFSS. Fig. 4 shows ZBfor three different corrugations from
which case (c) shows the most favorable response. For an
efficient LWA, the real part of ZBshould match with the
TL characteristic impedance Z0and the reactive part of ZB
should be almost zero within the desired frequency range. This
signifies favorable matching across the operating frequency
region at balanced condition. The proposed design shows
better matching as compared to exponentially or uniformly
oriented slot corrugations. When the balanced condition was
attained, left-handed and right-handed contributions exactly
balanced each other at a particular frequency. Also, the loss
profile is quite stable within the full operating band resulting
desired narrow beamwidth (∼9◦) with almost better than 15
dB improved side lobe level. The obtained balancing procedure
of the unit cell is described next through correlating the
full-wave simulation and corresponding circuit analysis. The
layout of the proposed unit cell is depicted in Fig. 6 with its
dimensions.
B. Equivalent Circuit and Dispersion Analysis
In Fig. 3(b), the achieved balanced condition of the proposed
unit cell was shown through dispersion analysis only based on
the full-wave simulations. The final results were validated by
comparing the full wave simulation obtained using the HFSS
with those obtained using the equivalent circuit analysis in
ADS. Actually, the equivalent circuit model in the present situ-
ation was initially predicted using the HFSS. The actual values
of the equivalent circuit elements are then determined using
the gradient optimization technique in ADS, by comparing the
S-parameters of the ADS with those of the HFSS. The detailed
methodology of the proposed design is depicted by a design
flow chart as shown in Fig. 5. It is clear from Fig. 5, the initial
values of the equivalent circuit are predicted from unbalanced
unit-cell S-parameter responses of HFSS by fitting the circuit
model in ADS. Further, the equivalent model parameters
are tuned in ADS to achieve balance condition. For circuit
analysis, half-sinusoidally varied IDC slot based HMSIW was
considered as a T-network comprising of RH (LR,CR) and LH
(LL,CL) elements. The detailed and precise equivalent circuit
is depicted in Fig. 7. The change in circuit element values
from the initial prediction gives an insight into the direction
of required change in physical parameters of the unit cell in
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4
Fig. 5. Design flow for the equivalent circuit of proposed design in ADS circuit simulator and correlation to HFSS design.
Fig. 6. Layout of the proposed unit cell with modified half-sinusoidally varied
IDC slots. Optimized dimensions were aHM SI W = 1.9 mm, dvia = 0.2 mm,
svia = 0.4 mm, wint = 0.14 mm, wg= 0.12 mm, lint = 0.22 mm, ∆1 =
0.22 mm, ∆2 = 0.16 mm, ∆3 = 0.2 mm and wp= 0.5 mm.
HFSS design to achieve balanced condition. For example, the
change in values of CLindicates the corresponding change
in gap between IDCs wg. The parallel inductors LLand
capacitors CRare coming into picture due to the metallic vias
considering the fringing field at the edges. The parameters LR
and CLappear due to the top-bottom metallic plane and the
IDCs. Consequently, the normalized phase constant β/k0of
the lossless CRLH unit cell was considered for our analysis
as (1)[20]
β=1
pcos−11−ω2LRCR−1
ω2LLCL
+CR
CL
+LR
LL(1)
The series and shunt cut-off frequencies, fse and fsh come due
to β= 0 for both the LH and RH dispersion curves. It is well
known that these frequencies are equal at balanced condition
of the unit cell as (2)[20]
fse =1
2π√LRCL
=fsh =1
2π√LLCR
(2)
Fig. 7. T-type equivalent circuit of the unit cell.
At this frequency point, the broadside radiation was obtained
with the inequality (3):
LR
LL
=CR
CL
(3)
The optimized circuit parameters are CR= 0.6 pF, LR= 0.35
nH, CL= 0.02 pF and LL= 0.011 nH. The dispersion curve
of the proposed geometry having 10 unit-cells can also be
calculated by computing the βand αparameters using the
following expressions (4) and (5)[51]:
β=1
p
Im cosh−11−S11S22 +S12 S21
2S21
(4)
α=1
p
Re cosh−11−S11S22 +S12 S21
2S21
(5)
After estimating the βand α, the dispersion diagram repre-
senting the factors |β|/k0and |α|/k0with k0=2πf /c as a
function of frequency fwas plotted as shown in Fig. 8 with the
same from equivalent circuit analysis. In our proposed design,
the desired balanced condition was achieved by appropriately
tuning the design parameters and it was obtained at 60 GHz.
The normalized phase constant |β|/k0ensures that the left side
of 60 GHz acts as LH media, sustaining upto 55 GHz, whereas
the right side acts as RH media with 65 GHz maximum. The
normalized attenuation constant |α|/k0exhibits very smooth
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Transactions on Antennas and Propagation
5
Fig. 8. Comparison of dispersion analysis between HFSS simulation and
equivalent circuit design in ADS.
Fig. 9. Realization of infinitely long leaky-wave antenna imposing periodic
boundary condition on a single unit-cell.
Fig. 10. Comparison of normalized Bloch impedance of the proposed antenna
for ideal case (infinitely long) and the designed prototype having 10 unit cells.
behavior with frequency, revealing frequency invariant shape
of the radiated beam.
C. Bloch Impedance Analysis
The Bloch impedance (ZB) is one of the key parameter
in designing leaky-wave antenna that indicates the impedance
matching of the geometry. Since the radiating discontinuity
due to perturbations is usually very small, the ZBcan be
realized as the input impedance of the infinite structure formed
by a cascading of the unit cells (Fig. 9). The value of ZB
varies within the unit cell, but it is periodically identical at
each periodic cross sectional plane and can be retrieved from
the field solution by simulation and also numerically in terms
of ABCD parameters as (6)[20]
ZB=VN
IN
=VN+1
IN+1
=2B
(D−A) + p(A+D)2−4(6)
where, VNand INare the voltage and current at input terminal
of Nth unit cell, respectively. Primarily, the above mentioned
equation was used for numerical estimation of ZBas an
ideal condition where the range of N were chosen from 0
to infinity. Basically, the Bloch impedance ZBis identical
at each terminal plane, hence during full-wave simulation, to
reduce the simulation time, the periodic boundary condition
was applied on the single unit cell to realize the infinite
periodic structure. Next, to validate the impedance matching
for the proposed design having N=10, ZBis retrieved from the
simulated S11 and S21 parameters of the finite structure with
10 unit cells. In Fig. 10, the normalized ZBof the infinitely
long LWA is compared with that of the finitely long LWA with
10 unit cells and it shows a good accordance to each other.
ZB=±Z0s(1 + S11)2−S2
21
(1 −S11)2−S2
21
(7)
D. Controlling Parameter of Antenna Leakage Rate
Since the HMSIW open magnetic wall contains maximum
electric field concentration, simple continuous corrugation
with half-sinusoidally varied amplitude has very limited con-
trol over leakage rate. Hence, the periodic gap wpwas kept
to provide better control over leakage and improved radiation
pattern. The change of wpimplied that how loosely or tightly
the IDCs of the unit cell were oriented within the leaky-wave
structure as shown in Fig. 12. Since, the variation of wpdidn’t
make impact on periodicity (p) and slot width (wint), hence,
the balanced condition was maintained for the unit cell but
at different frequencies. Also, the leakage rate in terms of
attenuation constant was changed accordingly, resulting into
beamwidth variation of the radiated main beam for different
frequencies. When the physical parameter wpwas comparable
with wg, the balanced frequency point shifted at 58 GHz and
leakage loss factor in terms of normalized attenuation constant
(αl/k0) increased with slight notch at balanced frequency as
shown in Fig. 11(a). As a result, the beamwidth increased
by 11◦since it is proportional with leakage loss factor as
shown in Fig. 11(d). Similar behavior of increased leakage loss
factor and beamwidth degradation were observed when wpwas
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6
Fig. 11. The variation of wpand its effect on (a)-(c) dispersion and (d)-(f) radiation characteristics.
Fig. 12. The variation of wpand its effect on antenna structure.
Fig. 13. Variation of radiator length and its effect on antenna gain.
Fig. 14. Variation of radiator length and its effect on remaining power at the
output port.
much larger than wgas shown in Fig. 11(c) and Fig. 11(f).
Hence, for achieving appropriate leakage rate from the leaky-
wave structure to maintain a desired narrow radiated beam and
broadside frequency at 60 GHz, wpwas optimized as 0.5 mm,
little higher than wg(0.12 mm) as shown in Fig. 11(b) and
Fig. 11(e).
III. V-BAND LE AK Y-WAVE ANT EN NA DE SI GN P ROC ES S
To realize the proposed leaky-wave antenna, the unit cell
was placed with periodicity palong the x-axis, providing
compactness and good beam steering performance. Fig. 1
shows the design of finite lossless structure which was opti-
mized by full wave simulation to demonstrate the effectiveness
of the concept. The half-power beamwidth and main lobe
angle were considered at the center frequency (60 GHz) as
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7
the design input parameters. The proposed antenna scanning
angle depends upon the nth spatial harmonic, βnand free
space wave number, k0. The antenna operated in the fast
wave region where the range of βnis -k0< βn<k0
having the backward (θ=-90◦) to forward quadrant (θ=90◦)
including broadside direction (θ=0◦) scanning capability. For
the uniform LWAs or quasi-uniform LWAs are realized by fast-
wave structures having a complex wavenumber kz=β−jα,
with the phase constant βin the range 0< β < k0. In
this case, the radiation usually takes place only in forward
direction using the fundamental radiating space harmonic.
However, to achieve the full space scanning like backward
quadrant to forward quadrant through broadside direction,
the composite right/left-handed (CRLH) media which is an
artificial or metamaterial transmission line that supports a fast
wave, should be implemented like the proposed antenna. The
structure, due to its metamaterial regime (p≪λg), operates in
the fundamental space harmonic, n=0 and supports backward
and forward wave propagation in the left-handed (LH) and
right-handed (RH) regions, respectively (-k0< β0< k0).
The direction of maximum radiation, θm, from z-axis can be
expressed as (8)[21]
θm= sin−1βn
k0= sin−1β0
k0
+nλ0
p(8)
After determining the βn, the total antenna length (Lant) was
also calculated based on (9)
∆θ≃λ0
Lant cos θ0≃
αl,total
k0
0.183.cos θm
(9)
where Lant,∆θ, and λ0are total antenna length, 3-dB
beamwidth, and free space wavelength, respectively. To
achieve narrow beam with compact geometry, the leakage
rate in terms of normalized attenuation constant must be
stabilized. The largest directivity of our proposed antenna was
occurred for the full antenna length of 46 mm with radiating
section length of 5.6λ0which contains 10 unit cells. It is
evident condition for highly efficient LWA that almost 90-
95%power should radiate from the structure and remaining
10-5%power should reach at the other port of the antenna
which is negligible amount to create reflection mismatch. To
determine the best case, the radiator length was gradually
increased like 5 unit cells (3.4λ0), 7 unit cells (4.3λ0), 10 unit
cells (5.6λ0), 15 unit cells (8.3λ0) and 20 unit cells (10.6λ0)
where the gain was increased accordingly as shown in Fig. 13.
The corresponding remaining power at the other port which
is a function of total attenuation constant and radiator length,
was also deeply studied which is shown in Fig. 14. From,
Fig. 13 and Fig. 14, it is clear that almost 97%power was
radiated from the structure for the radiator length 5.6λ0with
peak gain variation from 14-17.23 dBi. Beyond 8.3λ0, the peak
gain was further increased, however, it didn’t show significant
improvement. In summary, the further increment of 10 unit
cells (5λ0) after 5.6λ0long radiator length i.e. doubled the
radiator length from the proposed length resulted only 1.37
dB gain increment. Due to this, the most compact radiator
length is chosen as 5.6λ0for maximum directivity and gain
which contains 10 unit cells. For this purpose, keeping the
Fig. 15. Fabricated prototype of the proposed V-band antenna.
fabrication tolerances in mind, the maximum value of wint
to lint ratio was considered less than or equal to 0.15. Also,
the periodic perturbation in a single corrugation is maintained
as less than or equal to ≤λ0/10 to ensure the single beam
operation. The antenna was fabricated on Rogers RT/duroid
5880 substrate with ǫrof 2.2, tanδof 0.0009 and height of
0.38 mm.
IV. RES ULT S AN D DIS CU SSION
Fig. 15 shows the designed antenna fabricated on a single
PCB for testing and to validate the proposed concept ex-
perimentally. The fabricated prototype was only 5.6λ0long,
incorporating 10 unit cells, each with 8 slot fingers. Fig.
16 compares simulated and measured normalized radiation
patterns at the xz-plane (H-plane) and yz-plane (E-plane). In
simulation, the maximum 3-dB beamwidth in H-plane was
achieved as 10.2◦at 56.5 GHz and for measurement, it was
around 9.15◦at 64.5 GHz. The measured 3-dB beamwidth
values were obtained as 8.124◦, 8.68◦, 9◦, 8.92◦, 8.75◦,
8.69◦and 9.15◦at 55, 56.5, 58, 60, 61.5, 63 and 64.5 GHz,
respectively. The simulated radiated fan beam scanned from -
72.5◦to 50◦whereas the measured beam covered -72◦to 48◦.
Broadside radiation occurred at 60 GHz and measured side-
lobe level exceeded 15 dB throughout the working band. It
is observed that the measured pointing angle for the radiated
beam slightly differs from the simulated patterns. Basically,
due to very thin substrate thickness of 0.38 mm as compared
to the attached heavy load of V-band wave launchers and
associated cables during pattern measurement, the substrate
bent a little and that probably the cause of the slight changes
of the beam pointing angle. Also, the simulated beamwidth
was found little broader than measured one upto 57.5 GHz
and this is because of the variation of total leakage loss factor
of the structure (Fig. 17) which is proportional to the 3 dB
beamwidth as shown in eq. (9). Fig. 18 shows S-parameter
responses for the proposed structure confirmed operation from
almost 55-65 GHz with 60 GHz center frequency, and -10 dB
reflection bandwidth of 16.66%.
Fig. 19 shows simulated and measured main beam direction
variance with excitation frequency matched well, with 3D
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Transactions on Antennas and Propagation
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Fig. 16. (a)-(g) H-plane Radiation patterns of the proposed antenna at different frequencies, (h) E-plane pattern at broadside frequency of 60 GHz.
TABLE I
PER FOR MA NCE CO MPAR ISO N OF PRO PO SED ST RUC TU RE WI TH CLO SE LY RELATE D V-BAN D LWAS
Reference Technology Freq. BW Radiator length GR(dBi) Scanning range η
[20] Multilayered LTCC with SIIG 58-67 GHz(14.4%) 5.62λ011.7 21◦(38◦to 7◦) 90%
[21] multilayered MEMS 52-63 GHz(17.3%) 3.25λ011.4 9◦(12◦to 3◦) -
[22] Bulky: IDW with 3D printing 50-75 GHz(40%) 8.33λ014.2 49◦(-9◦to +40◦) 75%
[23] Metasurface loaded luneburg lens 60 GHz - 28.26 (sim) 36◦90%
[24] PRS loaded on PCB 62-65 GHz(4.72%) 10.66λ013.66 (sim) 12◦(sim) 95.5%(sim)
[49] PCB based 57-66 GHz(14.63%) 16λ017(sim) 40◦(-28◦to +15◦) 87%(sim)
Proposed Simple single-layered PCB 55-65 GHz(16.66%) 5.6λ014.5 120◦(-72◦to +48◦) 88.2%
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Transactions on Antennas and Propagation
9
Fig. 17. Comparison of simulated and measured total leakage loss of the
proposed antenna.
50.0 52.5 55.0 57.5 60.0 62.5 65.0
-60
-50
-40
-30
-20
-10
0
|S
11
| |S
21
|
Simulated lossless case
|S
11
| |S
21
| Simulated lossy case
|S
11
| |S
21
|Measured case
S-Par ameters (dB)
Frequency (GHz)
Fig. 18. S-parameter responses of the proposed antenna.
radiation patterns shown in inset. The realized gain was ob-
tained by incorporating some mismatching losses and losses in
efficiency with the power gain of the antenna to yield its value
as shown in Fig. 20 along with the simulated directivity. Fig.
20 shows that simulated realized gain (GR) varied from 13.5-
16.5 dBi, whereas the measured gain was from 12-14.5 dBi.
To meet the desired gain requirement for commercial purpose,
the structure for 4 sections array was simulated and the gain
was increased upto 27 dBi with compact dimension of 11λ0
×3.54λ0. In the presented paper, the proposed beam scanning
leaky-wave antenna geometry is a prototype that may fulfill
the desired specifications of the commercial antennas for the
mentioned V-band communication applications. Finally, the
radiation efficiency (η) is calculated by dividing the measured
peak gain by the simulated directivity and it varied from 50%
to 71%as depicted in Fig. 20. The measurement setup during
radiation pattern measurement in standard anechoic chamber
with V-band horn and test antenna is shown in Fig. 22.
Table I compares the proposed antenna with previously
proposed V-band leaky-wave antenna structures. References,
[20] and [21] had the similar frequency region of interest
and comparable overall antenna dimensions, but the proposed
geometry produced significantly improved gain and scanning
range. Reference [22] antenna exhibits comparable gain with
Fig. 19. Variations of beam direction with frequencies.
55.0 57.5 60.0 62.5 65.0
3
6
9
12
15
18
21
Gain/D irectivit y (dB)
Frequency (GHz)
Simulated realized gain
Measured realized gain
Simulated directivity
0
20
40
60
80
Radiation efficiency
Radiat ion effic iency (% )
Fig. 20. Variation of realized gain, directivity and radiation efficiency of the
proposed antenna.
Fig. 21. Measurement set-up in anechoic chamber during radiation pattern
measurements.
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10
much larger bandwidth, but the proposed structure is advan-
tageous in terms of smaller radiator size with significantly
large scanning range and radiation efficiency. References [24]
and [49] proposed PCB based designs, but the responses
were unsatisfactory and also the detailed measurement was
not performed. Hence, with the help of above comparison
table, one can infer that the proposed prototype of radiating
structure appears to be a better alternative as compared with
previous reported designs to facilitate wider scanning range,
higher gain, and larger radiation efficiency with more compact
geometry.
V. CONCLUSI ON
This paper presents a novel approach in designing com-
pact and planar V-band leaky-wave antenna. The proposed
design employs a HMSIW in association with modified half-
sinusoidally varied periodic slots which led to a compact
effective high performance antenna. The proposed V-band
leaky-wave antenna can operate over 55-65 GHz with the
main radiated beam continuously scanning from -72◦to 48◦
while retaining reasonably compact geometry (5.6λ0). The
maximum measured realized gain was obtained of 14.5 dBi
at 60 GHz. Specific periodic corrugated slot placement not
only significantly increased bandwidth and scanning range,
but also improved cross-polar level and gain. The proposed
prototype was designed using simple standard PCB fabrication
processes, providing cost effectiveness and bulk production
opportunities for industry fabrication. Performance of the pro-
posed prototype showed considerable advantages over recently
proposed V-band LWAs regarding scanning range, size, gain,
and radiation efficiency. Thus the proposed approach would
be a potential candidate for several mm-wave applications.
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