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A Broadband Circularly-Polarized Single-Layer
Metasurface Antenna Using Characteristic Mode
Analysis
Ahmed El Yousfi, Abdenasser Lamkaddem, Kerlos Atia Abdalmalak, Senior Member, IEEE, and Daniel
Segovia-Vargas., Senior Member, IEEE
Abstract—In this work, we propose a low-profile single-layer
coplanar waveguide (CPW)-fed metasurface (MTS) antenna with
broadband circular polarization radiation. With the help of
characteristic mode analysis (CMA), a 3 ×4 metasurface is
analyzed to reveal the useful modes supported by the structure.
Consequently, two modes with orthogonal current distribution,
broadside radiation, and nearly 90º phase difference over a wide
frequency band are chosen as operation modes. Moreover, the
modal near-field of the aforementioned modes shows that, unlike
conventional microstrip patches, the entire proposed metasurface
supports two kinds of extraordinary TM modes namely e- TM30
and e-TM04. Accordingly, a rotated CPW feeding line is used
to excite the two modes without adding an extra layer as
reported in the literature, making the design simpler and easier
to manufacture. Based on that, a low profile antenna of 0.058λ0
has been designed and fabricated. The measured results show an
impedance bandwidth (IBW) of 25% (4.87-6.26 GHz), 3-dB axial
ratio band (AR) of 19.42% (5.30-6.44 GHz), and a maximum gain
of 8 dBi.
Index Terms—Broadband, circular polarization, characteristic
mode, single-layer, metasurface.
I. INTRODUCTION
WITH the rapid development of wireless communica-
tion systems, antennas with circular polarization (CP)
features have received more and more attention due to their
advantages over linearly polarized (LP) ones such as the
orientation independence of the transmitter and receiver, and
their immunity to multipath distortion [1], [2].
One of the simplest methods to obtain circular polarization
is to use a single-feed microstrip antenna [3]-[5]. However, the
resultant bandwidth in terms of either a 3-dB axial ratio or -10
dB reflection coefficient is narrower (typically of 1.5% in the
AR band) because of only exciting one mode. The use of a
coplanar waveguide feeding is another simple technique used
in microstrip antennas [6]-[8]. Despite their easy simple layer
structure, most previous antennas are either linearly polarized
or have a narrow AR bandwidth for CP designs [7]. To increase
the AR bandwidth many approaches have been proposed. The
multi-feeding method is a commonly used technique for wide-
band performances [9], [10]. Compared with the single-feed
This work was supported by PID2019-109984RB-C41.
A. El Yousfi, A. Lamkaddem, and D.S.Vargas are with Signal Theory and
Communication Department, University Carlos III of Madrid, 28911 Madrid,
Spain (email: ahmed, abdenasser, dani@tsc.uc3m.es).
K.A. Abdalmalak is with the Signal Theory and Communication Depart-
ment, University Carlos III of Madrid, 28911 Madrid, Spain, and Electrical
Engineering Department, Aswan University, Aswan 81542, Egypt (e-mail:
kerlos@tsc.uc3m.es).
method, this approach requires a complex feeding network.
Therefore, much interest is dedicated to improving the single-
feed impedance and AR bandwidths designs. Alternatively,
multi-mode excitation single-feed designs have been proposed
to improve the AR bandwidth [11], [12]. For example, in [10],
a patch antenna loaded with a set of shorted pins results in
a significant increase in the AR band under the excitation
of triple mode resonances. Even though the bandwidth is
improved, it is not enough for current applications since it
is only 5.5% and has relatively high complexity due to the
presence of shorting pins. Similarly, three modes namely
TM10, TM01 , and a slot mode have been identified and excited
in a U-shaped slot patch, and TM10, TM01 , and TM11 in an
E-shaped slot patch [12]. As a result, a 21% wideband CP
performance is achieved. However, the main drawback of the
previous designs is its increased profile with a height of about
0.115λ0. Using multilayer structures and parasitic elements is
another solution for enhancing the AR bandwidth as reported
in [13], [14].
During the last years, the use of metasurfaces and meta-
materials has emerged as a widely used way to improve
antenna performance [15]- [19]. The metasurface is usually
placed above the radiating element either with or without an
air gap loading it and modifying its performance. In [15] a
metasurface is used above a patch antenna to convert linear to
circular polarization waves. The 3-dB axial ratio band reaches
8.1% and the impedance band rises to 17%. Another 4×4V-
shaped metasurface has been proposed to improve the axial
ratio band [18] achieving an AR of 15.95% with a wide
impedance bandwidth of 33%. Although the band is enhanced,
these antennas suffer from somewhat mechanical fragility due
to the presence of the air gap, relatively high profile, and a
relatively low 3-dB axial ratio band. Several designs consisting
of stacking the radiating element and the metasurface together
and removing the air gap have been proposed to overcome
the previous difficulties [20]- [23]. Although metasurfaces
can effectively broaden the band, their low gain and high
profile limit their application. Simpler designs based on single-
layer MTS antennas have been proposed in [24]- [27]. The
approach used in these antennas is mainly based on designing
a single antenna (generally a patch) that is initially capable of
radiating CP waves, then the MTS is used to improve IBW and
AR bandwidths or gain performances. However, the resultant
bandwidths are still narrower than typically 12%.
Other way, characteristic mode analysis (CMA) has been
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
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2
extensively used in the last two decades for designing and
analyzing antennas, despite being proposed in 1960 [28]-
[30]. The CMA provides physical insights that allow for
improving the antenna performances by either reducing cross-
polarization and increasing the port isolation [31], enhancing
bandwidth [32]-[34], or achieving omnidirectional metasurface
antenna [35], [36], or circular polarization [20], [22], [12].
The application of CMA for broadband circular polarization
MTS antenna has not been so common but its potential
can be exploited. For instance, in [41] a non-uniform low-
profile, broadband MTS CP antenna using CMA has been
proposed. Despite the low profile of the radiating element,
the main drawback of the design is the use of four substrate
layers with several shorting pins for the feeding network that
increase the complexity of the proposed antenna. To the best
of our knowledge, very few MTS antennas have been proposed
to achieve broadband circular polarization radiation with a
single layer and low profile. The only CPW single-layer MTS
antennas were presented in [8], [44]. However, the proposed
antennas are either linearly polarized [8] or circularly polarized
but at the expense of a very narrow 3-dB axial ratio bandwidth
[44].
In this paper, a broadband circularly polarized single-layer
metasurface antenna is proposed. A 3×4metasurface is
first studied using characteristic mode analysis without the
feeding line to select the useful modes. Therefore two modes
with orthogonal current have the potential to achieve circular
polarization radiation. Then a rotated CPW feed line is imple-
mented to simultaneously excite both modes. The rest of the
paper is organized as follows: Section II presents the study
of the metasurface through characteristic mode analysis, then
an appropriate feeding line is proposed to excite the structure.
The simulation and experimental results, which agree well,
are given in section III. Finally, a conclusion is presented in
section IV
II. PROPOSED ANTENNA DESIGN
The proposed single-layer metasurface antenna is illustrated
in Fig 1. The top part consists of 3×4rectangular patch unit
cell, whereas the bottom side contains the coplanar waveguide
feeding line (CPW) with a rotation angle α(Fig. 1(c)). Both
two parts are on a thin substrate of Rogers RT5880 with a
relative permittivity of 2.2 and a loss tangent of 0.002.
By properly adjusting the rotated angle αa good circular
polarization performance can be obtained. We note also that
linear polarization, left, and right-handed circular polarizations
can be obtained by setting the value of αequal to 0, +α, and
−αrespectively.
A. Analysis of 3×4metasurface without feeding line using
characteristic mode analysis
First, we start our design by undertaking the characteristic
mode analysis in CST of a 3×4metasurface without a
feeding line in the 4-8 GHz frequency band. It must be
noted that the dielectric and the ground plane are considered
infinite in the x-y plane in the simulation of characteristic
modes. The CMA provides two important parameters that are
Fig. 1. Proposed single-layer antenna (a) rectangular patch (b)3×4 metasur-
face (c) CPW feeding line (d) side view.
(a) (b)
Fig. 2. Modal significance of the proposed metasurface (a) phase difference
between mode 1 and mode 2 (b).
useful for designing a CP antenna: modal significance MS and
characteristic angle αn[29],[30]. A mode is resonant when
MS = 1 whereas when M S = 0 the mode is non-resonant.
To get CP radiation, at least two orthogonal modes should
be excited simultaneously with equal MS (or comparable),
characteristic angle difference of 90º, and same directivity in
the desired direction.
The modal significance of the first four modes is presented
in Fig. 2 (a). Mode 1 resonates around 5.5 GHz and has
large bandwidth ranging from 5 to 6.7 GHz (the bandwidth
is determined according to the criteria that MS ≥0.707). As
the frequency increases other modes are involved in the band
and have a significant contribution to the radiation if they are
properly excited. Mode 2 has resonance in the vicinity of 6.8
GHz with a wideband performance (5.9-7.3 GHz). Mode 3
resonates at 5.8 GHz with a relatively narrow band (5.6- 6.4
GHz) compared with the previous ones. Finally, mode 4 has
also a broadband modal significance (5.9- 7.4 GHz) with a
resonance frequency of 6.5 GHz. It is important to note that we
have only shown four modes because the other ones resonate at
higher frequencies and their modal far-field radiation patterns
have a null in the +z direction which is not useful in our case
[37].
To completely characterize the radiation performance of the
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
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3
(a) (b) (c) (d)
Fig. 3. Modal current distribution of the first four modes at 5.8 GHz (a)
mode 1 (b) mode 2 (c) mode 3 (d) mode 4.
(a) (b) (c) (d)
Fig. 4. Modal radiation pattern of the first four modes at 5.8 GHz (a) mode
1 (b) mode 2 (c) mode 3 (d) mode 4.
(a) (b)
(c)
Fig. 5. Variation of the phase difference of modes 1 and 2 for (a) length Lp
(b) width length Wp(c) gap length g.
(a) (b)
Fig. 6. Modal electric field top (perspective view) bottom (side view) at 5.8
GHz (a) mode 1 (b) mode 2.
(a) (b) (c)
(d) (e) (f)
Fig. 7. Modal current distribution for mode 1 at (a) 5.2 GHz (b) 5.4 GHz
(c) 5.6 GHz and mode 2 at (d) 5.2 GHz (e) 5.4 GHz (f) 5.6 GHz.
(a) (b)
Fig. 8. Modal magnetic field at 5.8 GHz (a) mode 1 (b) mode 2.
metasurface, the modal current distribution, and the modal far-
field of the first four modes are shown in Fig. 3 and Fig. 4
respectively. It is seen that mode 1 and mode 2 have an in-
phase current directed along the x and y-axis respectively. The
two other modes are out of phase, therefore their contribution
to the broadside far-field radiation is null. Fig. 4 presents the
modal far-field of the first four modes where modes 1 and 2
present a broadside radiation pattern whereas modes 3 and 4
have a null in the +z direction. These results are consistent
with the modal current shown in Fig. 3. Usually to achieve
circular polarization a phase difference of about 90º between
the selected modes is required. Thus, the phase difference
between modes 1 and 2 as a function of frequency is illustrated
in Fig. 2 (b). It is seen that a phase difference of more than 60º
is obtained over a wide frequency band ranging from 5.5 to
6.3 GHz. Although this difference is not exactly 90º, a proper
choice of a well-designed feeding line can compensate for it
as demonstrated in [20]. Moreover, it is observed that at 5.8
GHz the phase difference between the two modes reaches a
peak of 70º while their corresponding MS is 0.9 for mode1
and 0.7 for mode 2 which are quite close in terms of MS. On
the other hand, at 6.3 GHz, Modes 1 and 2 have equal MS
with a phase difference of nearly 60º. These two frequencies
correspond to the two dips obtained in the AR plot (AR=1.2
dB at 5.8 GHz, AR=0.45 dB at 6.3 GHz) as will be shown in
Fig. 14(b).
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
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4
Fig. 9. Modal weighting coefficient (MWC) of the first four modes of the
MTS antenna.
As mentioned, an optimal phase difference of 90º of the
selected modes is needed for CP radiation [21]. Therefore,
a parametric study is provided to evaluate the influence of
length Lp, width Wp, and the gap g between the adjacent
patch unit cells on the phase difference of modes 1 and 2(see
Fig. 5). Apparently, the phase difference between mode 1 and
mode 2 can be easily tuned by the parameters of the MTS
unit cell. More specifically, the phase difference increases with
the increase of Lpwhereas Wphas an adverse effect as the
increase of Wp results in a decrease in the phase difference.
The gap g has little influence on the phase difference as clearly
seen in Fig. 5 (c). Therefore, a good compromise between all
the parameters is needed to achieve optimal results.
To gain additional physical insights into the whole MTS, the
modal electric near field is investigated along with the current
distribution. As seen in Fig. 3, since mode 1 and mode 2 have
an in-phase current along the x and y-axis respectively, they
can be considered sort of extraordinary TM modes: e-TM30
and e-TM04 respectively. This can be clearly seen from the E-
field distribution of both modes as shown in Fig. 6. A TM10
and TM01 can be found underneath each sub-patch, therefore
resulting in extraordinary TM modes for the entire MTS (three
nulls along the x-axis and four nulls in the y-axis for mode 1
and 2 respectively ). We note that the main difference between
TM modes (TM30 and TM04) supported by a conventional
patch is that the currents along the patch are out of phase
giving rise to side lobes [17].
It is well known that CMA depends on the frequency [36]
which may result in variation in the modal current and modal
radiation pattern. Therefore the current distribution of modes
1 and 2 at different frequencies is illustrated in Fig. 7. From
the figure, it can be seen that both modes have a stable modal
current distribution over the frequency band with a maximum
current located in the center cells of the metasurface (denoted
by a circle in Fig. 7). This result guarantees a broadside
radiation pattern of the metasurface over a wide frequency
range and indicates the location of the feeding line.
B. Metasurface with feeding line
Based on the modal analysis done in the previous section, a
rotated coplanar waveguide is proposed for exciting modes 1
(a) (b)
Fig. 10. Simulated current distribution at 5.8 GHz for different phases of the
input signal (a) ϕ= 0º (b) ϕ= 90º.
(a) (b)
Fig. 11. Effect of rotated angle αon (a) S11 (b) AR.
and 2 as shown in Fig. 1(c). The feeding is a magnetic current
source corresponding to a slot. Furthermore, for a slot, vi
nis
given as
vi
n=⟨Hn, M ⟩=ZZS
Hn.M ds. (1)
where Hnis the modal magnetic field of the nth mode and
Mis the magnetic current on the slot. Thus, to efficiently
excite modes 1 and 2 the feeding slot should be located at
the position where the modal current is strong, in addition,
Hnand Mshould be parallel. From Fig. 8 it is observed that
the intensity of the modal magnetic field of modes 1 and 2
(a) (b)
Fig. 12. Effect of feeding length Lf1on (a) S11 (b) AR.
(a) (b)
Fig. 13. Effect of number of metasurface elements on (a) S11 (b) AR.
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
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5
is strong around the center unit cells along the x and y-axis.
In this design, the rotated slot is placed along the diagonal of
the substrate to provide two magnetic current components Mx
and Myalong the x and y-axis respectively.
The feeding is selected so that two electric field components
with orthogonal polarizations associated with the aperture can
excite the two modes. From Fig. 3 and Fig. 7, it is noticed that
the maximum current is located in the central patches of the
metasurface however the corners cell has a weak current distri-
bution. Therefore the feeding line should be placed underneath
the center of the metasurface. Besides this kind of feeding
offers a low profile and simple structure as it contains only one
layer. The modal weighting coefficient (MWC), which shows
into which modes the most power is coupled in the presence
of the feeding line [37], is calculated by placing an equivalent
infinitesimal magnetic dipole surrogate along the diagonal of
MTS to emulate the CPW feeding line [42]. With respect to
Fig. 9, one can observe that only modes 1 and 2 are excited
within the interested band.
Finally, the simulated current distribution at 5.8 GHz for
different phases of the excitation input signal in the presence of
the CPW feed line is demonstrated in Fig. 10. For ϕ= 0º (Fig.
10(a)), the current is mainly oriented along the y-axis which
is very consistent with the current distribution associated with
the mode 2 (see Fig. 3(b)). Likewise, for ϕ= 90º (Fig. 10(b)),
the current is mostly directed along the x-axis which is similar
to that corresponding to mode 1 as seen in Fig. 3(a).
C. Optimization and parametric study
To get further insights into manipulating the frequency band
and for optimum performances, a parametric study of key
parameters is given in this section. The effect of the angle αon
the AR, and S11 is presented in Fig. 11. It is observed in Fig.
11 (a) that as αincreases the impedance bandwidth increases
with good matching at both lower and higher frequency points.
From Fig. 11 (b), it is seen that as αdecreases the first dip
in the AR goes down to lower frequencies. The second dip
at higher frequency (around 6.4 GHz) remains unchanged.
It should be mentioned that for α= 0 the MTS antenna is
linearly polarized as only mode 1 of the MTS is excited.
Another parameter of the feeding line that greatly affects
the impedance and 3-dB AR bandwidths is the length of the
rotated CPW line Lf1as illustrated in Fig. 12. It is seen
that a good impedance and 3-dB AR bandwidths can be
obtained by adjusting Lf1to 12 mm. The CPW slot width
Wfsolely affects the matching while its influence on AR can
be neglected.
The influence of the number of elements of metasurface on
the antenna performance is investigated in Fig. 13. From Fig.
13 (a), it is observed that the number of elements has much
influence on the upper frequency around 6.25 GHz, whereas
its effect on the lower one is weak. Decreasing the number
of elements shifts the upper frequency to the higher band,
resulting in a wide impedance bandwidth. Fig. 13 (b) shows
that when the number of elements increases the two dips in AR
move toward each other and merge resulting in a narrow AR
band. Therefore the optimum number of elements is selected
TABLE I
OPT IMI ZE D VALUE S OF TH E PRO PO SED A NTE NNA
Param. L W WpLpg LfWfLf1gfh
Value [mm] 55 55 9 12.2 0.5 27.5 1 12 0.2 3.2
(a) (b)
Fig. 14. Performance comparison of the 3×4 metasurface and the conventional
patch (a) reflection coefficient (b) axial ratio.
to be 3×4. It is important to note that the length Lpand width
Wpof the MTS unit cell can be adjusted to control the AR
and IBW (this is not shown for brevity).
The optimized dimensions of the proposed antenna design
are presented in Table I.
For a fair comparison, a conventional rectangular patch
antenna having the same dimensions as the total ones of
the MTS has been designed (see Fig. 1) and simulated. The
performance of the proposed MTS antenna is shown in Fig.
14 and compared with those of the rectangular patch antenna.
We can observe that with the same dimensions of a patch, a
broadband characteristic is obtained both in terms of 3-dB AR
and impedance bandwidths. It can also be mentioned that the
results of the CPW- based single-feed patch antenna are very
consistent with those in [7].
Fig. 15 shows the phase and magnitude differences of
the Exand Eycomponents in the far field along the +z
axis. It is seen that both phase and amplitude differences
cross the 90º line and 0 amplitude line at two points which
shows the presence of two minimum poles in the AR curve.
Moreover, the variation within the working band is smooth
which indicates a broad AR band.
Fig. 15. Magnitude and phase difference of Exand Eyin far-field in +z
axis.
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
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6
(a) (b) (c)
Fig. 16. Prototype of the proposed antenna (a) top view (b) back view (c)
far-field measurements set up.
Finally, some guidelines of the CPW feed metasurface
antenna design are summarized as follows:
Step 1: perform CMA of the metasurface and select the
potential modes to contribute to the CP radiation (section II-
A) and optimize the phase difference between the selected
modes (as shown in Fig. 5).
Step 2: chose the adequate type of feeding line and its
location based on the current distribution of the selected modes
(Section II-B).
Step 3: select the appropriate feeding line angle to obtain
the desired polarization radiation (α= 0 for LP, or αfor
LHCP/RHCP ).
Step 4: optimize the whole MTS antenna including the
feeding line to obtain better results in the targeted frequency
band (Lf1, length and width of the metasurface unit cell, and
slightly change α).
III. SIMULATION AND EXPERIMENTAL RESULTS
To verify the performance of the simulated results a pro-
totype of the proposed antenna is fabricated and measured as
shown in Fig. 16. The simulated and measured S11 and 3-
dB AR are compared in Fig. 17. The measured impedance
bandwidth and AR bands are 25.08% (4.868-6.264 GHz)
and 19.4%(5.30-6.43GHz) respectively. A good agreement
between simulation and measurement is seen. The simulated
and measured broadside gain along with simulated radiation
efficiency over the frequency is shown in Fig. 18. A flat gain
within the operating band with a maximum measured gain of
8 dBi is achieved. A good agreement is obtained in the whole
band except for a slight difference at lower frequencies (from
5 to 5.4 GHz). In addition, the proposed antenna has a good
radiation efficiency of more than 96% over the operating band.
The simulated and measured radiation patterns at 5.8, 6, and
6.2 Hz in the x-z and y-z planes of the proposed antenna
design are shown in Fig. 19. There is a good agreement
between simulated and measured results. The antenna has left-
handed circular polarization radiation. Moreover, The cross-
polarization level is less than -20 dB at the broadside direction
within the working band.
Table II presents the comparison of the proposed antenna
performances over some recent works. It is seen that our
proposed design shows good features in terms of 3- dB axial
ratio band and impedance bandwidth. Compared with designs
of two layers [15],[18],[38], [43] the proposed antenna has a
large impedance and 3-dB AR bands. Besides these designs
(a) (b)
Fig. 17. Simulated and measured (a) S11 (b) Axial ratio.
(a) (b)
Fig. 18. Simulated and measured (a) broadside gain (b) simulated radiation
efficiency.
(a) (b)
(c) (d)
(e) (f)
Fig. 19. Measured and simulated radiation pattern of the proposed antenna
in the x-z plane (left), and y-z plane (right)at 5.8 GHz (a)-(b),6 GHz (c)-(d),
and 6.2 GHz (e)-(f).
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: UNIVERSIDAD CARLOS III MADRID. Downloaded on January 30,2023 at 10:51:52 UTC from IEEE Xplore. Restrictions apply.
7
TABLE II
COMPARISON WITH RECENT WORKS
Ref. year. Num. of layers -10dB band [%] 3-dB AR[%] Peak Gain [dBi] Height (λ0) Size λ0xλ0Air gap
[43] 2022 2 25 14.7 8.6 0.137 diameter:1.3 Yes
[41] 2020 4 16.7 17.4 6.6 0.068 0.7*0.7 No
[15] 2013 2 17 8.1 10 0.067 0.98*0.98 Yes
[18] 2019 2 33 15.95 5.76 0.18 0.37*0.37 Yes
[38] 2019 2 15.7 13 10 0.085 NA Yes
[39] 2020 2 33.6 19.6 10.2 0.076 0.92*0.92 Yes
[20] 2018 2 38.5 14.3 9.4 0.072 1.4*1.4 No
[21] 2019 2 22 8.5 6.5 0.043 0.58*0.58 No
[22] 2021 2 28.2 20.9 9.7 0.07 1*1 No
[23] 2016 2 33.7 16.7 5.8 0.07 0.6*0.49 No
[40] 2020 2 19 11.4 7 0.06 0.78*0.78 No
[24] 2020 1 14.7 14.7 9.1 0.05 1.18*1.18 No
[25] 2020 1 23.4 16.8 11.3 0.04 1*1 No
[26] 2018 1 18 12.8 8.3 0.038 0.85*0.85 No
[27] 2017 1 19.5 12.9 9.8 0.028 0.92*0.92 No
[12] 2022 1 28.1 21 7.4 0.115 1.1*1.1 No
[12] 2022 1 25.8 25.4 7.3 0.114 1.1*1.1 No
Our work 1 25.08 19.42 8 0.058 1*1 No
Main features where the proposed antenna outperforms the reported ones are highlighted in orange.
contain an air gap between the metasurface and the antenna
which makes the profile high and increases the mechanical
problems. In the case of designs without air gaps, [20] and [41]
present a wide 3-dB AR band and impedance band. However,
using two or four layers increases the profile and makes the
fabrication process a bit complex. Finally, compared with
antenna designs consisting of only one printed layer [24], [25],
[26], and [27], our single-layer metasurface antenna shows
wideband in terms of both impedance and 3-dB axial ratio
bands. Although designs in [12] show a wideband CP radiation
compared to the proposed antenna at the cost of bulky volume.
IV. CONCLUSION
A single-layer, low-profile, and broadband circularly polar-
ized metasurface antenna has been presented. The investigation
of the modal behavior of the metasurface shows that two
modes with orthogonal current, almost 90º phase difference,
and broadside radiation pattern have the potential for CP
radiation. By splitting the patch into periodic 3×4metasurface,
two extraordinary TM modes (e-TM30 and e-TM04) have been
created which gives more physical insights into the whole
MTS. Based on this analysis, a rotated coplanar waveguide
feed line is chosen for excitation. RHCP/LHCP or linear
polarization can be achieved by properly adjusting the feeding
line rotation angle α. Finally, a fabricated prototype has been
measured to verify the simulations. A good agreement between
simulation and experimental results is obtained. The proposed
design antenna shows good performance which makes it a
good candidate for wireless communication systems.
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This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: UNIVERSIDAD CARLOS III MADRID. Downloaded on January 30,2023 at 10:51:52 UTC from IEEE Xplore. Restrictions apply.
8
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Ahmed El Yousfi was born in Al Hoceima, Mo-
rocco. He received the double bachelor’s degree
in physics from the University of Lille 1 Sciences
and Technology, France, and Mohamed I University,
Oujda, Morocco, in 2016, and the master’s degree
in electronics and telecommunications engineering
from the Abdelmalek Essaˆ
adi University of T´
etouan,
Morocco, in 2017. He is currently pursuing the Ph.D.
degree with the Group of Radiofrequency, Electro-
magnetics, Microwaves, and Antennas (GREMA),
Carlos III University of Madrid (UC3M). In 2020,
he joined the Signal Theory and Communication Department, UC3M, as
a Teaching Assistant. He worked on massive MIMO antennas for Huawei
Project, and Ferrite antennas for Indra. In 2022, he was a visiting Ph.D.
student in the Department of Electrical and Information Technology at Lund
University, Sweden.
He has authored/co-authored several international conference papers and
journal papers. He served as a peer reviewer in the IEEE Access journal. He
also served as a Peer Reviewer for the European Conference on Antennas and
Propagation (EuCAP).
His research interests include multiband/broadband antennas based on
metamaterials, characteristic mode analysis for metasurface antennas, array
antennas for 5G applications, and implantable antennas. He received an
Erasmus+ Grant, in 2019.
Abdenasser Lamkaddem was born in El Aioun
Sidi Mellouk, Morocco, in 1993. He received the
B. S. degree from Mohamed I University, Oujda,
Morocco, in 2014, and the M. S. degree in Com-
munication Systems and Embedded Electronics from
Abdelmalek Essaadi University, Tanger, Morocco, in
2016. He is currently pursuing the Ph.D. degree at
Carlos III University, Madrid, Spain.
In 2019, he was a Visiting Student in the De-
partment of Signal Theory and Communications,
Madrid, Spain. In 2023, he was a Visiting Research
Student in the Electrical and Electronic Engineering Department at the
University of Liverpool. He has participated in several research projects
financed by Telnet and Huawei.
He has authored/co-authored several international conference papers and
journal papers. He served as a peer reviewer in the IEEE Access and the
International Journal of Communication Systems (IJCS).
His research interests include implantable, wearable antennas and wireless
power transfer for biomedical applications, small antennas, antenna array
for 5G applications, Ultra-wideband antennas, reconfigurable antennas, EBG,
frequency selective surfaces, and Characteristic Mode Analysis. He received
the Erasmus+ grant in 2019.
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: UNIVERSIDAD CARLOS III MADRID. Downloaded on January 30,2023 at 10:51:52 UTC from IEEE Xplore. Restrictions apply.
9
Kerlos Atia Abdalmalak (S’19–M’21–SM’22) was
born in Luxor, Egypt in 1990. received the B.Sc.
degree (Hons.) (ranked 2nd among the colleagues) in
telecommunication engineering from Aswan Univer-
sity, Egypt, in 2011, and the M.Sc. degree in multi-
media and communication from Universidad Carlos
III de Madrid (UC3M), Spain in 2015, with an ex-
cellent grade. He received his Ph.D. from UC3M in
March 2022 in the field of antennas and radiometers
for radio astronomy applications (with an excellent
grade and cum laude/international mentions). He
joined the Department of Electrical Engineering at Aswan University as a
Teaching Assistant from 2011 to 2014. He worked as a Research Fellow
in GREMA research group (Radiofrequency, Electromagnetism, Microwaves
and Antennas Group) at UC3M, Spain from 2015 to 2022 and being a short-
term Visiting Scholar at Southern Methodist University (SMU), USA in 2018.
Currently, he is a Postdoc Researcher at Universidad Polit´
ecnica de Madrid
(UPM), Spain. He has authored/coauthored 60 reviewed papers published in
indexed journals and international conferences including 11 first-quartile (Q1)
Journal Citation Ranking (JCR) journals. He has participated in 15 research
projects financed by the Madrid Regional Ministry of Education, Ministry of
Economy and Business, Huawei, European Space Agency (ESA), SENER,
and other private companies with a total fund exceeding 3 million Euros. His
technical interests include antennas and propagation, ultrawideband/multiband
antennas, reflector/feed systems, radio astronomy receivers, mobile base sta-
tions, 5G MIMO communications, satellite remote sensing, Earth observation
radiometers, photonic nonlinear up-conversion, antenna arrays, metasurfaces
antennas, mm-wave/THz technologies, microwave/optical measurements, and
whispering gallery mode resonators. Mr. Abdalmalak served as a Guest
editor for Crystals and as a reviewer for several JCR journals such as
IEEE Transactions on Antennas and Propagation, Optics Express, IEEE
Access, Progress in Electromagnetics Research (PIER), Materials, Sensors,
Photonics, Applied Sciences, Physica Scripta, International Journal of Infrared
and Millimeter Waves (IJIM). Also served as a reviewer, technical program
committee (TPC), and publication chair for many international conferences
such as EuCAP, ITCE, CGMIP, AsiaSim, ICEECC, and CIAP. He received the
Erasmus Mundus GreenIT grant in 2014, the European School of Antennas
(ESoA) registration fee grant in 2015, and the Young Scientists Award (2nd
prize) by URSI/Spain in 2017. Was selected as IEEE Ambassador for the
IEEEXtreme 14.0 competition at Region 8 (Europe, Middle East, and Africa)
in 2020 and as the IEEE Section Lead for Spain for the IEEEXtreme 15.0
Competition in 2021. He received the outstanding Ph.D. Thesis Award by
Universidad Carlos III de Madrid (UC3M) in 2022, the best Ph.D. thesis in
the Aerospace field by Ayuntamiento de Madrid in 2022, and Margarita Salas
postdoc scholarship in 2022.
Daniel Segovia Vargas (Senior Member, IEEE) was
born in April 1968. He received the Telecommuni-
cations Engineering degree from ETSIT, UPM, in
1993, the Ph.D. degree (cum laude) in telecommuni-
cations engineering from ETSIT-UPM, in 1998, with
a distinction by unanimity, and the Doctor Honoris
Causa degree from Universidad Cat´
olica San Pablo,
Arequipa. From 1993 to 1998, he was an Assistant
Professor at the Universidad de Valladolid. Since
1998, he has been a Professor at the Universidad
Carlos III de Madrid (UC3M). Since 2001, he has
been an Associate Professor (Tenure) of signal theory and communications
at the signals at UC3M, where he is currently teaching high-frequency
microwave and circuits and antennas. Since 2003, he has been chairing
the Radiofrequency, Electromagnetics, Microwaves, and Antennas Group
(GREMA), UC3M. From 2004 to 2010, he was the Head of telecommu-
nications engineering at the Escuela Polit´
ecnica Superior, UC3M. From 2012
to November 2020, he was the Head of Escuela Polit´
ecnica Superior, UC3M.
He has been a Full Professor at UC3M, since 2016. He has been a Visiting
Researcher at the Rutherford Appleton Laboratory and CTU in Prague. He
has authored or coauthored more than 350 publications in scientific journals
and international conferences (more than 90 in indexed international journals
and more than 20 international invited conferences). His research interests
include antennas (antenna arrays and miniaturized antennas, where he has
led different projects with outstanding companies, such as Airbus, Repsol, or
Indra), active antennas, metamaterials, and technologies in THz frequencies.
He has been a member of AP-S Society, since 1998, and MTT-S Society, since
2001. He has been the Treasurer of the European Microwave Conference, in
2018, and Eucap 2022 in Madrid. He received the Best Thesis Award in
mobile communication by COIT-Ericsson for his Ph.D. degree. He was the
Chairperson of URSI2011 and a member of the Organizing Committee of
Eucap 2010 (where he was the Awards Committee). He has organized several
international workshops in the domain of metamaterials and THz technologies.
He has been chairing courses in the European School of Antennas, since 2013.
He has been the National Delegate for European Cost actions in the antennas
field (Cost 284, Cost IC0603, and Cost IC1102), since 2002. He has chaired
more than 80 research and development projects, both public and private.
Since 2013, he has been a Treasurer and a Secretary of the IEEE Spanish
Chapter.
This article has been accepted for publication in IEEE Transactions on Antennas and Propagation. This is the author's version which has not been fully edited and
content may change prior to final publication. Citation information: DOI 10.1109/TAP.2023.3239104
© 2023 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See https://www.ieee.org/publications/rights/index.html for more information.
Authorized licensed use limited to: UNIVERSIDAD CARLOS III MADRID. Downloaded on January 30,2023 at 10:51:52 UTC from IEEE Xplore. Restrictions apply.
