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(a) An illustration of the antenna-element model, (b) planar cut of the antenna-element model with a box highlighting the coaxial antenna feed, and simplified illustrations of the two different interfaces between the antenna coaxial feed (inside the box) and the PCB: (c) the direct interface and (d) the indirect interface. In the interface illustrations, the antenna metal is colored orange, copper on the PCB yellow, and the PCB substrate gray.
Source publication
The first electrically steerable dual-polarized Ka-band Vivaldi antenna array with connector-less, surface-mount interface is developed. The antenna interface includes integrated coaxial feed lines within the antenna element structure and contact pads on the PCB. The antenna array covers the frequency range from 26 to 40 GHz with a beam steering ra...
Contexts in source publication
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... design further and demonstrate electrical beam-steering using element-specific phase shifters. The antenna-PCB interface is redesigned to allow solder-less surface mounting of the antenna on the PCB. The novel connection is realized through capacitive coupling between the coaxial feeds of the antenna and the feeding pads on the PCB, as shown in Fig. 1. Furthermore, the PCB pads and antenna transition is designed to allow reasonable manufacturing and antenna alignment tolerance without performance degradation. When the metallic antenna is mounted directly on top of the PCB, the antenna structure could additionally be used to dissipate some of the heat produced by the electronic ...
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... interface can be implemented, for example, with the two different ways illustrated in Fig. 1 (c) and ...
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... side of the PCB and then to another layer through vias. The indirect transition can result in a simpler PCB design since distributing the transition through the PCB to the desired places without striplines inside them is easier. On the other hand, cavities are necessary in the antenna structure corresponding to the microstrip lines, as shown in Fig. ...
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... phase of the E-field across each of the antenna elements is shown in Fig. 10. The simulated results in the first column with identical signal in each antenna port results in a consistent phase across the antenna elements. The second column represent the measured results in which larger variation is present. In the first simulation, the phase across the antenna aperture varies less than 20 • between any two ...
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... are very close to the designed dimensions when inspected under a microscope. When the simulation is correlated with the data from the measurement, the simulation results agree well with the measurements, and similar phase patterns are observed. A comparison between the measured and simulated radiation patterns with correlated signals is shown in Fig. 11, exhibiting reasonable agreement. The radiation patterns of the initial simulation with ideal feed signals is shown for reference. The measured pattern differs mostly from the ideal simulation on the E-plane due to the flaw in the feed lines for the first and last rows. The characterization of the losses in the feed network is a lot ...
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... antenna array withouth the feeding network is also challenging if the loss in the feeding network is unknown. We use a compromise where the simulated loss of the feeding network is used to compensate for the network loss to estimate the gain of the array. The simulated gain of the antenna array and the estimation of the measured gain is shown in Fig. 12. The gain estimate is calculated by compensating the simulated feed network loss (8.5-14.7 dB) and using a SGH as a gain reference. The simulated results are performed with the bare antenna, where each element is fed directly from the coaxial feed line. The first simulation represents the ideal performance of the designed 8×8 array. In ...
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... is used as the calibration value. This value is calculated for each array element and is then used as the initial state value for each phase shifter. When the antenna array is electrically steered these values are combined with the progressive phase shift. The phase of the E-field in front of the antenna elements after the calibration is shown in Fig. 13 at 32 and 37 GHz for the broadside radiation pattern. The variation in the measured phases in front of the elements is now less than 40 • . The effect of the calibration can be seen in the Fig. 14 where measured radiation patterns with different steering angles are presented in E-, H-, and D-planes at 32 and 37 GHz. When comparing the ...
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... these values are combined with the progressive phase shift. The phase of the E-field in front of the antenna elements after the calibration is shown in Fig. 13 at 32 and 37 GHz for the broadside radiation pattern. The variation in the measured phases in front of the elements is now less than 40 • . The effect of the calibration can be seen in the Fig. 14 where measured radiation patterns with different steering angles are presented in E-, H-, and D-planes at 32 and 37 GHz. When comparing the broadside radiation pattern in Fig. 14 and Fig. 11 the broadside patterns especially at 37 GHz in the E-plane can be observed. In the uncalibrated case, the first and second side lobes are merged ...
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... the broadside radiation pattern. The variation in the measured phases in front of the elements is now less than 40 • . The effect of the calibration can be seen in the Fig. 14 where measured radiation patterns with different steering angles are presented in E-, H-, and D-planes at 32 and 37 GHz. When comparing the broadside radiation pattern in Fig. 14 and Fig. 11 the broadside patterns especially at 37 GHz in the E-plane can be observed. In the uncalibrated case, the first and second side lobes are merged together. However, in the calibrated case the side lobes are properly separated and the pattern is closer to the ideal simulated ...
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... envelopes are normalized to the broadside direction of the same plot at each frequency point. The cross-polarized data is represented as a relative value such that it is normalized to the co-polarized field in the same direction. The simulated co-polarized and cross-polarized envelopes of the 8×8 antenna array at 26 and 40 GHz are shown in Fig. 15 and exhibit consistent behavior at both edges of the frequency band. Simulations corresponding to the measurement results at 32, 34, 35, and 37 GHz are shown in Fig. 16. The results follow well the unit-cell simulations, i.e., the cosine pattern. The dashed line in the co-polarized envelope represents the −3 dB scan loss. In all the ...
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... it is normalized to the co-polarized field in the same direction. The simulated co-polarized and cross-polarized envelopes of the 8×8 antenna array at 26 and 40 GHz are shown in Fig. 15 and exhibit consistent behavior at both edges of the frequency band. Simulations corresponding to the measurement results at 32, 34, 35, and 37 GHz are shown in Fig. 16. The results follow well the unit-cell simulations, i.e., the cosine pattern. The dashed line in the co-polarized envelope represents the −3 dB scan loss. In all the cases the steering range with less than 3 dB of scan loss in E-and H-planes extends close to or beyond the 60 • circle. In the diagonal planes, the scan loss is higher by ...
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... measured envelopes of the co-and cross-polarized fields are shown in Fig. 17. The antenna array behaves similarly in the measurements as in the simulations. The shape of the −3-dB contour line in the co-polarized envelope is similar to the simulations and can be seen to extend near the 60 • circle in the E-and H-planes. The measured crosspolarized envelope pattern also displays a pattern similar to the ...
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... the measurements are limited to the operation range of the phase shifters, the simulations of the same 8×8-array have been simulated from 26 GHz to 40 GHz, and they are in good agreement with the unit-cell simulations. The simulations between 26 and 32 GHz and between 37 and 40 GHz exhibit similar behavior, as already presented in Fig. 15. The co-polarized scan loss is close to 3 dB at steering angle 60 • in the E-and H-planes whereas the level of the cross-polarization is high on the edges of the steering range away from the E-and ...
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Citations
... Some methods, such as using separate connectors [4], [20], waveguides [9], [21], and transitions between microstrip (MS) lines and waveguides [22], [23], [24] are unsuitable for the integrated mm-wave phased-array system because of the bulky size and only one polarization support. Even though managed to feed the separate dual-polarized antenna array as a surface-mount component [5], [17], [25], the method is complex and limited to antennas with coaxial ports. Therefore, an effective connection between the circuits on the PCB and the QRSW of the EQA remains an open question. ...
An evanescent quad-ridge antenna (EQA) fed by a compact transition is proposed in this letter for 5G millimeter-wave (mm-wave) phased-array applications. The EQA is realized by an evanescent-mode quad-ridged square waveguide (QRSW) filter with a cross-shaped slot as the radiator. The compact transition is designed by leveraging the coupling between circuits on a multi-layer printed circuit board (PCB) and metal ridges in the QRSW. Thanks to the proposed microstrip-to-QRSW transition, the EQA can be easily installed on a multi-layer PCB and fed by active circuits. The proposed antenna is fabricated and measured. Measurement results show that the antenna can cover the 5G n257 band from 26.5 GHz to 29.5 GHz. For beamforming demonstration, a 4 × 4 EQA array driven by mm-wave active beamformer ICs (BFICs) is designed and measured. The proposed EQA achieves compact size, filtering function, and easy integration with active circuits, which makes it suitable for 5G mm-wave phased-array applications.
... Considerable research efforts have been made to tackle these design challenges of mmWave 5G phased arrays [7][8][9][10][11][12][13][14][15][16][17][18][19][20][21], summarized and compared in Table 1. However, most works have focused on only one or two performance parameters. ...
... To evaluate the compactness of the array, this research introduced a density metric for the array. This metric is calculated as the inverse relationship between the total [7] Sequential rotation-fed dual-polarized patch 0.028 [8] Probe-fed patch antenna 0.037 [9] Parasitically gap-coupled patch 0.042 [10] PCB-integrated stacked-patch antenna 0.051 [11] Tightly coupled dipole array 0.083 [12,13] Slotted waveguide array 0.083 [14] Air-cavity-backed patch antenna on lid substrate 0.114 [15] Cavity-backed patch antenna 0.186 [16] Dielectric resonator antenna arrays 0.198 [17] Surface-mounted Vivaldi array 0.222 [18] Coaxial-fed stacked patch antenna 0.259 [19] Single-polarized surface-mounted Vivaldi array 0.265 [20] Coaxial-fed stacked patch antenna 0.434 This work Stacked patch with notched corners, EBG, and element rotation 0.451 volume of the array and the same element numbers of half wavelength cubed as in (7). In the end, the number of polarizations (Pol) is also considered in the FoM to characterize the polarization performance. ...
Millimeter (mmWave) 5G phased arrays require multiple simultaneous features for reliable high data-rate communication. However, it is challenging to simultaneously achieve a true wideband operation for all parameters due to mutual coupling and grating lobe issues. A 5 W 5-stacked patch rectangular phased array was designed and fabricated in a 15-layer low-temperature co-firing ceramic (LTCC) substrate. This work utilized multiple design strategies, such as employing stacked patch topology, electromagnetic band gap (EBG) structures, and the rotation of elements to obtain a true wideband performance. The single element of the phased array was a dual linear polarized stacked patch antenna with notched corners. Compared to a standard patch antenna, the bandwidth was enhanced by 15.3%. The undesired mutual coupling between elements was minimized by rotating nearby elements and introducing EBG structures between the adjacent elements. A wideband beamforming network composed of a Rotman lens and a 5-way Wilkinson power divider (WPD) was also designed and fabricated. The proposed phased array achieved 6 GHz of bandwidth, covering 24 to 30 GHz and achieving a maximum gain of 17.5 dBi and a wide beam-scanning range from –50 to +50 degrees. This work also introduced a figure of merit (FoM) based on all critical performance parameters for objective comparison with state-of-the-art designs. The proposed design achieved the highest FoM (0.451), whereas most similar 5G phased array designs achieved much lower FoM value.
... In this paper, 28 GHz has been selected because of fabrication considerations. Many types of antenna are used in Millimeter wave band such as Fermi, Vivaldi, and Quasi Yagi [4][5][6]. ...
In this paper, a Millimeter Wave antenna is designed which has a compact, lightweight and planar configuration. The frequency band is 27–29.5 GHz as a candidate band for 5G/millimeter Wave (mmW) systems. The antenna structure is similar to Quasi Yagi antenna which has driver, director, feeding part, and reflector. Substrate Integrated Waveguide used for feeding part of single element antenna and Wilkinson power divider used for antenna array feeding network. The proposed antenna has been simulated, fabricated, and measured (S11, E, and H pattern). The simulation and measurement values showed good similarity. The switched line phase shifter used to consider the beam steering (rotation) ability of the designed antenna which is important in (mmW) systems such as RADARs and mobile handsets. To evaluate this ability, for 0°, 30°, 45°, and 90° phase differences, the beam steering angle (θ) simulated and also for −90° and 90° implemented. The results showed that the efficiency, S11, and E patterns in the rotated beam is suitable and without degradation in antenna operation. To simulate and evaluate the designed antenna HFSS and CST Software are used.
... The reason for this limitation is that the operating bandwidth of TSAs can be increased primarily by longer flare elements exacerbating the imbalance between vertical and horizontal current potentials [207]. Notably in [187], a surface-mounted Vivaldi antenna array for the entire Ka-band has been presented with a scanning range of ±60°in the principal and ±50°in the diagonal planes, respectively. ...
Satellite, fifth generation (5G), and sixth generation (6G) mobile communication systems operating at micro- and millimeter wave frequencies are an essential pillar in advanced network architectures for high-throughput low latency services. The decisive factors for this current development were the significant technological advances accomplished over the last two decades, which meanwhile enable highly integrated, feature-rich, and cost-effective realizations of complete phased array transceiver topologies. Motivated through the today's trend for heterogeneous constellation types to provide truly global coverage, this contribution reviews the current state-of-the-art of electronically steerable antennas for terrestrial and non-terrestrial communication systems up to 100 GHz. First, the potential benefits and limitations of the most relevant technologies are contrasted and put into context with recent system architectures for adaptive beamforming. Their operating principles along with various experimental implementation and achievements providing advanced capabilities such as multi-band/multi-beam operation, polarization agility, and wide-angle scanning are thoroughly presented. Particular emphasis is laid on the review of direct radiating arrays, quasi-optical antenna configurations, and metasurface-based antennas.
... In this paper, the objective is to extend the design presented in [27]- [29] and realize the envisioned integration capability of these structures. In the previous works, a similar type of antenna element design is used to confirm the feasibility of the surface-mounted antenna design and the applicability of additive manufacturing to the design to retain good performance up to 40 GHz. ...
... An antenna array prototype is designed integrating previously presented ideas [27]- [29]. The core idea of the antenna array is to be wideband, beam-steerable, modular, and to theoretically enable the array to be extended to any size and shape. ...
... The antenna element is a single piece of metal and can be manufactured using additive manufacturing or electric discharge machining. It is based on the design presented in [27]- [29] with the dimensions scaled and optimized for the frequency range up to 30 GHz. One antenna element is dual polarized, and the elements can be used to construct arrays of various sizes. ...
A modular dual-polarized Vivaldi antenna array design for 18–30 GHz frequency is presented. The array module consists of the antenna and RF modules. The antenna module comprises
$4\times 4$
dual-polarized antenna elements with element spacing of
$\lambda $
/2 at 30 GHz. The RF module contains the amplifiers and phase shifters that control all the elements using commercial off-the-shelf (COTS) flip-chip components. The footprint of the RF module is the same as that of the antenna module allowing assembly of antenna arrays of almost any size and shape. Additionally, the interface between the antenna and the RF modules is connectorless, decreasing the number of components required in the assembly and decreasing the overall cost of the system. An array of
$4\times 8$
dual-polarized antenna elements is constructed from two array modules to prove the seamless operation of the modular design. The prototype uses Anokiwave chips with a frequency range from 26.5 GHz to 29.5 GHz. The measured amplitude and phase of the electric field in front of the antenna aperture is uniform so as to equally feed the elements. Additionally, the demonstrated beam steering up to ±60° in the plane of the larger array dimension matches well with the simulations, proving the feasibility of the design.
... Adding parasitic patches [14], [15], dielectric lens [16], [17], and metamaterials [18], [19] could be useful for increasing the antenna gain and improving the antenna pattern. Consequently, Vivaldi antennas have been widely applied for wireless communications [20], array creations [21], and microwave imaging applications [22]. Not many Vivaldi antennas have been developed to possess CP property [23] - [27]. ...
This paper proposes a novel ultrawideband antenna having high gain and circular polarization. Its circular polarization can be easily changed to right handed or left handed. The antenna consists of four sophisticated antipodal Vivaldi antennas and an ultrawideband feeding network. The feeding network composes of four horizontal-to-vertical transitions, a two-to-four power divider and a 3-dB directional coupler. This network can provide two pairs of outputs with equal magnitude and ninety-degree phase difference. In addition, it is horizontally fed with a single input port which can be directly connected to the corresponding circuit board. The newly introduced transition can lessen highly the discontinuity between the horizontally placed circuit and the vertically stood antenna. A prototype antenna is designed, fabricated, and tested. Its impedance-matching, circularly-polarized and stable-gain band is sufficiently wide to cover the entire Ku, K, and Ka bands. The fractional bandwidth (FBW) is as large as 113.2%. In the whole band, the antenna axial ratio is well below 3 dB while the antenna gain is stable and up to 18.35 dBic. The remarkable performance is the first time achieved in the literature.
... In this paper, we extend the research on the Vivaldi antenna array design presented in [5], [12] by demonstrating the use of AM processes, namely, SLM and binder jetting to manufacture the fully metallic antenna array. While AM has been demonstrated in horn antenna elements even above 200 GHz [11], it has yet to be demonstrated on more complex antenna array geometries at mm-wave frequencies. ...
... consequently the high density of the antenna feeds, it is extremely difficult to use separate RF-connectors and cables for each antenna element. Instead of feeding each antenna element directly with a cable, the antenna array is surface mountable, and the elements are fed with air-filled coaxial transmission lines which are integrated in the structure and pass through the antenna ground plane [12]. The coaxial transmission lines in the antenna structure couple capacitively to circular pads on the PCB shown in Fig. 3 (a). ...
... In this demonstration, where the manufacturing methods are compared, a fixed feed network for one polarization is used for smaller amount of uncertainties. A more thorough analysis of the feed network with beam steering is presented in [12]. ...
Additive manufacturing (AM) is a rapidly developing field which potentially decreases the manufacturing costs and enables increasingly complex antenna shapes. Metal-based AM might be particularly useful to manufacture antennas at mm-wave range, because there antennas are physically small enough making additive manufacturing cost efficient, and manufacturing accuracy could still suffice for good electrical performance. In this paper, two additively manufactured and an identical machined fully metallic Ka-band Vivaldi antenna arrays are compared. The manufactured antenna arrays are compared using RF-measurements to conclude the feasibility of AM for manufacturing antenna arrays at mm-wave frequencies. Comparison of the measured radiation patterns and realized gains of each of the antenna arrays between 26 and 40 GHz shows close to identical radiation patterns for all the arrays. A loss in efficiency of 0.51.5 dB is observed in the AM arrays when compared to the machined array due to the used materials and the surface roughness.
This paper comprises a fractal Koch curve with tapered slot structure subjected to magnetic field strengthanalysis for Ultra High Frequencies (UHF) RFID near field reader antenna. In the design, Variable distance with opposite direction current principle is used. The center frequency of the proposed antenna operates at 897 MHz with bandwidth of 101 MHZ. (853 to 954 MHz). The obtained bandwidth is sufficient to cover the UHF RFID frequency for, Canada (902-928 MHz), Europe andUSA (865-868 MHz). The near field analysis was verified by keeping different liquids and the antenna performance is calculated. During the measurement, the antenna is kept at a distance of 50 mm in z direction and is observed that the interrogation area of proposed antenna is achieved with 243 mm X 190 mm. The result shows that the antenna exhibits the ability to track chemicals in industries and pharmaceutical applicationsmanagement. It is also observed that the proposed antenna is sufficient to operate as commercial tags for near field magnetic induction communication.
